LTC3805EMSE-5#TRPBF - LINEAR TECHNOLOGY

LTC3805-5

38055fe

FEATURES DESCRIPTION

Adjustable Frequency

Current Mode Flyback/

Boost/SEPIC DC/DC Controller

The LTC

3805-5 is a current mode DC/DC controller de-

signed to drive an N-channel MOSFET in ﬂyback, boost

and SEPIC converter applications. Operating frequency

and slope compensation can be programmed by external

resistors. Programmable overcurrent sensing protects

the converter from overload and short-circuit conditions.

Soft-start can be programmed using an external capacitor

and the soft-start capacitor also programs an automatic

restart feature.

The LTC3805-5 provides ±1.5% output voltage accuracy

and consumes only 360µA of quiescent current during

normal operation and only 40µA during micropower start-

up. Using a 9.5V internal shunt regulator, the LTC3805-5

can be powered from a high VIN through a resistor or it

can be powered directly from a low impedance DC voltage

from 4.7V to 8.8V.

The LTC3805-5 is available in the 10-lead MSOP package

and the 3mm × 3mm DFN package.

L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered

trademarks, No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners.

5V to 12V/1A Boost Converter

TYPICAL APPLICATION

n VIN and VOUT Limited Only by External Components

n 4.5V Undervoltage Lockout Threshold

n Adjustable Slope Compensation

n Adjustable Overcurrent Protection with Automatic

Restart

n Adjustable Operating Frequency (70kHz to 700kHz)

With One External Resistor

n Synchronizable to an External Clock

n ±1.5% Reference Accuracy

n Only 100mV Current Sense Voltage Drop

n RUN Pin with Precision Threshold and Adjustable

Hysteresis

n Programmable Soft-Start with One External Capacitor

n Low Quiescent Current: 360µA

n Small 10-Lead MSOP and 3mm × 3mm DFN

APPLICATIONS

n Automotive Power Supplies

n Telecom Power Supplies

n Isolated Electronic Equipment

n Auxiliary/Housekeeping Power Supplies

n Power over Ethernet

Efﬁciency and Power Loss vs Load Current

LTC3805-5

GND

ITH

RUN GATE

ISENSE

SSFLT

FS FB

VCC

118k

8mΩ

22µF

×2

SYNC

1.33k

4.3µH UPS840

VIN

VOUT

12V

13.7k

191k

100µF

38055 TA01a

0.1µF

20k

470pF

LOAD CURRENT (A)

0.01

EFFICIENCY (%)

POWER LOSS (W)

38055TA01b

75 0.1 110

100

1.5

0.5

2.5

2.0

5V EFFICIENCY

5V LOSS

8.5V LOSS

8.5V EFFICIENCY

LTC3805-5

38055fe

ABSOLUTE MAXIMUM RATINGS

VCC to GND

Low Impedance Source ........................ –0.3V to 8.8V

Current Fed ........................................25mA into VCC*

SYNC ........................................................... –0.3V to 6V

SSFLT........................................................... –0.3V to 5V

FB, ITH, FS ................................................. –0.3V to 3.5V

RUN ........................................................... –0.3V to 18V

(Note 1)

OC, ISENSE .................................................... –0.3V to 1V

Operating Junction Temperature Range

(Notes 2, 3) ........................................... –55°C to 150°C

Storage Temperature Range .................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)

MSE Package .................................................... 300°C

*LTC3805-5 internal clamp circuit regulates VCC voltage to 9.5V

TOP VIEW

DD PACKAGE

10-LEAD (3mm × 3mm) PLASTIC DFN

1GATE

VCC

ISENSE

SYNC

SSFLT

ITH

RUN

TJMAX = 125°C, θJA = 45°C/W

EXPOSED PAD (PIN 11) IS GND, MUST BE CONNECTED TO GND

SSFLT

ITH

RUN

GATE

VCC

ISENSE

SYNC

TOP VIEW

MSE PACKAGE

10-LEAD PLASTIC MSOP

TJMAX = 125°C, θJA = 45°C/W

EXPOSED PAD (PIN 11) IS GND, MUST BE CONNECTED TO GND

PIN CONFIGURATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC3805EMSE-5#PBF LTC3805EMSE-5#TRPBF LTDGX 10-Lead Plastic MSOP –40°C to 85°C

LTC3805IMSE-5#PBF LTC3805IMSE-5#TRPBF LTDGX 10-Lead Plastic MSOP –40°C to 125°C

LTC3805HMSE-5#PBF LTC3805HMSE-5#TRPBF LTDGX 10-Lead Plastic MSOP –40°C to 150°C

LTC3805MPMSE-5#PBF LTC3805MPMSE-5#TRPBF LTDGX 10-Lead Plastic MSOP –55°C to 150°C

LTC3805EDD-5#PBF LTC3805EDD-5#TRPBF LDHB 10-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C

LTC3805IDD-5#PBF LTC3805IDD-5#TRPBF LDHB 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C

Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges. *The temperature grade is identiﬁed by a label on the shipping container.

Consult LTC Marketing for information on lead based ﬁnish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/

ORDER INFORMATION

LTC3805-5

38055fe

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

VTURNON VCC Turn-On Voltage VCC Rising l4.3 4.5 4.7 V

VTURNOFF VCC Turn-Off Voltage VCC Falling l3.75 3.95 4.15 V

VHYST VCC Hysteresis 0.55 V

VCLAMP1mA VCC Shunt Regulator Voltage ICC = 1mA, VRUN = 0 l8.8 9.25 9.65 V

VCLAMP25mA VCC Shunt Regulator Voltage ICC = 25mA, VRUN = 0 l8.9 9.5 9.9 V

ICC Input DC Supply Current Normal Operation (fOSC = 200kHz)

(Note 4)

360 µA

VRUN < VRUNON or VCC < VTURNON – 100mV

(Micropower Start-Up)

l40 90 µA

VRUNON RUN Turn-On Voltage VCC = VTURNON + 100mV l1.122 1.207 1.292 V

VRUNOFF RUN Turn-Off Voltage VCC = VTURNON + 100mV l1.092 1.170 1.248 V

IRUN(HYST) RUN Hysteresis Current l4 5 5.8 µA

VFB Regulated Feedback Voltage 0°C ≤ TJ ≤ 85°C (E-Grade) (Note 5)

–40°C ≤ TJ ≤ 85°C (E-Grade) (Note 5)

–40°C ≤ TJ ≤ 125°C (I-Grade) (Note 5)

–40°C ≤ TJ ≤ 150°C (H-Grade) (Note 5)

–55°C ≤ TJ ≤ 150°C (MP-Grade)

0.788

0.780

0.770

0.800

0.812

0.820

IFB VFB Input Current VITH = 1.3V (Note 5) 20 nA

gmError Ampliﬁer Transconductance ITH Pin Load = ±5µA (Note 5) 333 µA/V

DVO(LINE) Output Voltage Line Regulation VTURNOFF < VCC < VCLAMP1mA (Note 5) 0.05 mV/V

DVO(LOAD) Output Voltage Load Regulation ITH Sinking 5µA (Note 5)

ITH Sourcing 5µA (Note 5)

mV/µA

fOSC Oscillator Frequency RFS = 350k 70 kHz

RFS = 36k 700 kHz

DCON(MIN) Minimum Switch-On Duty Cycle fOSC = 200kHz 6 9 %

DCON(MAX) Maximum Switch-On Duty Cycle fOSC = 200kHz 70 80 95 %

fSYNC As a Function of fOSC 70kHz < fOSC < 700kHz,

70kHz < fSYNC < 700kHz

67 133 %

VSYNC Minimum SYNC Amplitude 2.9 V

ISS Soft-Start Current –6 µA

IFTO Fault Timeout Current 2 µA

tSS(INT) Internal Soft-Start Time No External Capacitor on SSFLT 1.8 ms

tFTO(INT) Internal Fault Timeout No External Capacitor on SSFLT 4.5 ms

tRISE Gate Drive Rise Time CLOAD = 3000pF 30 ns

tFALL Gate Drive Fall Time CLOAD = 3000pF 30 ns

VI(MAX) Peak Current Sense Voltage RSL = 0 (Note 6) l85 100 115 mV

ISL(MAX) Peak Slope Compensation Output Current (Note 7) 10 µA

VOCT Overcurrent Threshold ROC = 0 (Note 8) l85 100 115 mV

IOC Overcurrent Threshold Adjust Current 10 µA

ELECTRICAL CHARACTERISTICS

The l denotes the speciﬁcations which apply over the speciﬁed operating

junction temperature range, otherwise speciﬁcations are at TA = 25°C, VCC = 5V, unless otherwise noted (Note 2).

LTC3805-5

38055fe

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The LTC3805-5 is tested under pulsed conditions such that

TJ ≈ TA. The LTC3805E-5 is guaranteed to meet speciﬁcations from

0°C to 85°C. Speciﬁcations over the –40°C to 85°C operating junction

temperature range are assured by design, characterization and correlation

with statistical process controls. The LTC3805I-5 is guaranteed over the

–40°C to 125°C operating junction temperature range, the LTC3805H-5 is

guaranteed over the full –40°C to 150°C operating junction temperature

range and the LTC3805MP-5 is tested and guaranteed over the full –55°C

to 150°C operating temperature range. High junction temperatures

degrade operating lifetimes. Operating lifetime is derated at junction

temperatures greater than 125°C. Note that the maximum ambient

temperature consistent with these speciﬁcations is determined by speciﬁc

operating conditions in conjunction with board layout, the rated package

thermal resistance and other environmental factors.

Note 3: TJ is calculated from the ambient temperature TA and power

dissipation PD according to the following formula:

TJ = TA + (PD • 45°C/W)

Note 4: Dynamic supply current is higher due to the gate charge being

delivered at the switching frequency.

Note 5: The LTC3805-5 is tested in a feedback loop that servos VFB to the

output of the error ampliﬁer while maintaining ITH at the midpoint of the

current limit range.

Note 6: Peak current sense voltage is reduced dependent on duty cycle

and an optional external resistor in series with the SENSE pin. For

details, refer to Programmable Slope Compensation in the Applications

Information section.

Note 7: Guaranteed by design.

Note 8: Overcurrent threshold voltage is reduced dependent on an

optional external resistor in series with the OC pin. For details, refer to

Programmable Overcurrent in the Applications Information section.

LTC3805-5

38055fe

TEMPERATURE (°C)

–75 –50

788

VFB VOLTAGE (mV)

792

800

808

796

804

812

–25 0 25 50

38055 G01

75 100 150125

TYPICAL PERFORMANCE CHARACTERISTICS

Reference Voltage

vs Temperature

Reference Voltage

vs Supply Voltage

Oscillator Frequency vs RFS

Oscillator Frequency

vs Supply Voltage

Oscillator Frequency

vs Temperature

RUN Undervoltage Lockout

Thresholds vs Temperature

RUN Hysteresis Current

vs Temperature

VCC Undervoltage Lockout

Thresholds vs Temperature

RFS (kΩ)

fOSC (kHz)

400

500

600

400

38055 G03

300

200

0100 200 300

100

800

700

VCC (V)

0.799700

VFB (V)

0.799800

0.799900

0.800000

0.800100

0.800200

0.800300

6 85 7 9 10

38055 G02

VCC (V)

201

202

203

38055 G04

200

199

5 6 7 9

198

197

196

fOSC (kHz)

RFS = 124kΩ

TEMPERATURE (°C)

198

OSCILLATOR FREQUENCY (kHz)

199

200

201

202

38055 G05

RFS = 124kΩ

–75 –50 –25 0 25 50 75 100 150125 –75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

1.170

RUN VOLTAGE (V)

1.175

1.185

1.195

1.180

1.190

1.200

1.205

38055 G06

VRUN(ON)

VRUN(OFF)

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

4.70

IRUN(HYST)

4.80

5.00

4.90

5.10

5.20

38055 G07

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

3.0

VCC UNDERVOLTAGE LOCKOUT (V)

3.5

4.5

4.0

5.0

38055 G08

VTURN(ON)

VTURN(OFF)

LTC3805-5

38055fe

Start-Up ICC Supply Current

vs Temperature

ICC Supply Current

vs Temperature

VCC Shunt Regulator Voltage

vs Temperature

Peak Current Sense Voltage

vs Temperature

Overcurrent Threshold

vs Temperature

Internal Soft-Start Time

vs Temperature

External Soft-Start Current

vs Temperature

External Timeout Current

vs Temperature

TYPICAL PERFORMANCE CHARACTERISTICS

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

START-UP SUPPLY CURRENT (µA)

38055 G09

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

300

SUPPLY CURRENT (µA)

310

330

320

340

360

350

38055 G10

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

9.20

VCLAMP (V)

9.25

9.35

9.45

9.30

9.40

9.50

9.55

9.60

38055 G11

VCLAMP25mA

VCLAMP1mA

TEMPERATURE (°C)

99.0

PEAK CURRENT SENSE VOLTAGE (mV)

99.2

99.6

100.0

99.4

99.8

100.2

100.4

100.8

100.6

38055 G12

–75 –50 –25 0 25 50 75 100 150125 –75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

OVERCURRENT THRESHOLD (mV)

100

102

104

38055 G13

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

1.90

INTERNAL SOFT-START TIME (ms)

1.95

2.05

2.15

2.00

2.10

2.20

38055 G14

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

4.4

4.5

ISS SOFT-START CURRENT (µA)

4.6

4.8

5.0

4.7

4.9

5.1

5.2

5.4

5.3

38055 G15

–75 –50 –25 0 25 50 75 100 150125

TEMPERATURE (°C)

1.5

IFTO EXTERNAL TIMEOUT CURRENT (µA)

1.6

1.8

2.0

1.7

1.9

2.1

38055 G16

LTC3805-5

38055fe

BLOCK DIAGRAM

PIN FUNCTIONS

SSFLT (Pin 1): Soft-Start Pin. A capacitor placed from

this pin to GND (Exposed Pad) controls the rate of rise of

converter output voltage during start-up. This capacitor is

also used for time out after a fault prior to restart.

ITH (Pin 2): Error Ampliﬁer Compensation Point. Normal

operating voltage range is clamped between 0.7V and

1.9V.

FB (Pin 3): Receives the feedback voltage from an external

resistor divider across the output.

RUN (Pin 4): An external resistor divider connects this pin

to VIN and sets the thresholds for converter operation.

FS (Pin 5): A resistor connected from this pin to ground

sets the frequency of operation.

SYNC (Pin 6): Input to synchronize the oscillator to an

external source.

ISENSE (Pin 7): Performs two functions: for current mode

control, it monitors the switch current, using the voltage

across an external current sense resistor. Pin 7 also injects

a current ramp that develops slope compensation voltage

across an optional external programming resistor.

OC (Pin 8): Overcurrent Pin. Connect this pin to the ex-

ternal switch current sense resistor. An additional resistor

programs the overcurrent trip level.

VCC (Pin 9): Supply Pin. A capacitor must closely decouple

VCC to GND (Exposed Pad).

GATE (Pin 10): Gate Drive for the External N-Channel

MOSFET. This pin swings from GND to VCC.

GND (Exposed Pad Pin 11): Ground. A capacitor must

closely decouple GND to VCC (Pin 9). The exposed pad

must be soldered to electrical ground on PCB for electrical

contact and rated thermal performance.

–

SLOPE

COMP

CURRENT

RAMP

VCC RUN

GATE

DRIVER GATE

ISENSE

38055 BD

OSCILLATOR

CURRENT

COMPARATOR

OVERCURRENT

COMPARATOR

SHUTDOWN

ITH CLAMPS

20mV

ITH

ERROR

AMPLIFIER

SOFT-START RAMP

10µA

100mV

SWITCHING

LOGIC AND

BLANKING

CIRCUIT

SSFLT

GND

1.2V

2SYNCFS

5 6

9 4

UNDERVOLTAGE

LOCKOUT

800mV

REFERENCE

SOFT-START

FAULT

LTC3805-5

38055fe

OPERATION

The LTC3805-5 is a programmable-frequency current mode

controller for ﬂyback, boost and SEPIC DC/DC converters.

The LTC3805-5 is designed so that none of its pins need

to come in contact with the input or output voltages of

the power supply circuit of which it is a part, allowing

the conversion of voltages well beyond the LTC3805-5’s

absolute maximum ratings.

Main Control Loop

Please refer to the Block Diagram of this data sheet and

the Typical Application shown on the front page. An ex-

ternal resistive voltage divider presents a fraction of the

output voltage to the FB pin. The divider is designed so

that when the output is at the desired voltage, the FB pin

voltage equals the 800mV internal reference voltage. If

the load current increases, the output voltage decreases

slightly, causing the FB pin voltage to fall below the 800mV

reference. The error ampliﬁer responds by feeding current

into the ITH pin causing its voltage to rise. Conversely, if

the load current decreases, the FB voltage rises above the

800mV reference and the error ampliﬁer sinks current away

from the ITH pin causing its voltage to fall.

The voltage at the ITH pin controls the pulse-width modula-

tor formed by the oscillator, current comparator and SR

latch. Speciﬁcally, the voltage at the ITH pin sets the cur-

rent comparator’s trip threshold. The current comparator’s

ISENSE input monitors the voltage across an external current

sense resistor in series with the source of the external

MOSFET. At the start of a cycle, the LTC3805-5’s oscillator

sets the SR latch and turns on the external power MOSFET.

The current through the external power MOSFET rises as

does the voltage on the ISENSE pin. The LTC3805-5’s cur-

rent comparator trips when the voltage on the ISENSE pin

exceeds a voltage proportional to the voltage on the ITH pin.

This resets the SR latch and turns off the external power

MOSFET. In this way, the peak current levels through the

external MOSFET and the ﬂyback transformer’s primary

and secondary windings are controlled by the voltage on

the ITH pin. If the current comparator does not trip, the

LTC3805-5 automatically limits the duty cycle to 80%,

resets the SR latch, and turns off the external MOSFET.

The path from the FB pin, through the error ampliﬁer,

current comparator and the SR latch implements the

closed-loop current mode control required to regulate the

output voltage against changes in input voltage or output

current. For example, if the load current increases, the

output voltage decreases slightly, and sensing this, the

error ampliﬁer sources current from the ITH pin, raising

the current comparator threshold, thus increasing the peak

currents through the transformer primary and secondary.

This delivers more current to the load and restores the

output voltage to the desired level.

The ITH pin serves as the compensation point for the control

loop. Typically, an external series RC network is connected

from ITH to ground and is chosen for optimal response to

load and line transients. The impedance of this RC network

converts the output current of the error ampliﬁer to the

ITH voltage which sets the current comparator threshold

and commands considerable inﬂuence over the dynamics

of the voltage regulation loop.

LTC3805-5

38055fe

Figure 1. Start-Up/Shutdown State Diagram

LTC3805-5

SHUTDOWN

LTC3805-5

FAULT

TIMEOUT

38055 F01

VRUN > VRUNON

AND VCC > VTURNON

VRUN < VRUNOFF

VCC < VTURNOFF

VSSFLT < 0.7V VOC > 100mV

LTC3805-5

ENABLED

OPERATION

Start-Up/Shutdown

The LTC3805-5 has two shutdown mechanisms to disable

and enable operation: an undervoltage lockout on the VCC

supply pin voltage, and a precision-threshold RUN pin. The

voltage on both pins must exceed the appropriate threshold

before operation is enabled. The LTC3805-5 transitions

into and out of shutdown according to the state diagram

shown in Figure 1. Operation in fault timeout is discussed

in a subsequent section. During shutdown the LTC3805-5

draws only a small 40µA current.

The undervoltage lockout (UVLO) mechanism prevents

the LTC3805-5 from trying to drive the external MOSFET

gate with insufﬁcient voltage on the VCC pin. The voltage

at the VCC pin must initially exceed VTURNON = 4.5V to en-

able LTC3805-5 operation. After operation is enabled, the

voltage on the VCC pin may fall as low as VTURNOFF = 4V

before undervoltage lockout disables the LTC3805-5. See

the Applications Information section for more detail.

The RUN pin is connected to the input voltage using a

voltage divider. Converter operation is enabled when the

voltage on the RUN pin exceeds VRUNON = 1.207V and

disabled when the voltage falls below VRUNOFF = 1.170V.

Additional hysteresis is added by a 5µA current source

acting on the voltage divider’s Thevenin resistance. Setting

the input voltage range and hysteresis is further discussed

in the Applications Information section.

Setting the Oscillator Frequency

Connect a frequency set resistor RFS from the FS pin to

ground to set the oscillator frequency over a range from

70kHz to 700kHz. The oscillator frequency is calculated

from:

fOSC =24 •109

RFS −1500

The oscillator may be synchronized to an external clock

using the SYNC input. The rising edge of the external clock

on the SYNC pin triggers the beginning of a switching

period, i.e., the GATE pin going high. The pulse width of

the external clock is quite ﬂexible.

Overcurrent Protection

With the OC pin connected to the external MOSFET’s current

sense resistor, the converter is protected in the event of

an overload or short-circuit on the output. During normal

operation the peak value of current in the external MOSFET,

as measured by the current sense resistor (plus any adjust-

ment for slope compensation), is set by the voltage on the

ITH pin operating through the current comparator. As the

output current increases, so does the voltage on the ITH

pin and so does the peak MOSFET current.

LTC3805-5

38055fe

OPERATION

First, consider operation without overcurrent protection.

For some maximum converter output current, the voltage

on the ITH pin rises to and is clamped at approximately 1.9V.

This corresponds to a 100mV limit on the voltage at the

ISENSE pin. As the output current is further increased, the

duty cycle is reduced as the output voltage sags. However,

the peak current in the external MOSFET is limited by the

100mV threshold at the ISENSE pin.

As the output current is increased further, eventually,

the duty cycle is reduced to the 6% minimum. Since the

external MOSFET is always turned on for this minimum

amount of time, the current comparator no longer limits

the current through the external MOSFET based on the

100mV threshold. If the output current continues to in-

crease, the current through the MOSFET could rise to a

level that would damage the converter.

To prevent damage, the overcurrent pin, OC, is also

connected to the current sense resistor, and a fault is

triggered if the voltage on the OC pin exceeds 100mV. To

protect itself, the converter stops operating as described

in the next section. External resistors can be used to ad-

just the overcurrent threshold to voltages higher or lower

than 100mV as described in the Applications Information

section.

Soft-Start and Fault Timeout Operation

The soft-start and fault timeout of the LTC3805-5 uses either

a ﬁxed internal timer or an external timer programmed

by a capacitor from the SSFLT pin to GND. The internal

soft-start and fault timeout times are minimums and can

be increased by placing a capacitor from the SSFLT pin

to GND. Operation is shown in Figure 1.

Leave the SSFLT pin open to use the internal soft-start and

fault timeout. The internal soft-start is complete in about

1.8ms. In the event of an overcurrent as detected by the

OC pin exceeding 100mV, the LTC3805-5 shuts down and

an internal timing circuit waits for a fault timeout of about

4.25ms and then restarts the converter.

Add a capacitor CSS from the SSFLT pin to GND to increase

both the soft-start time and the time for fault timeout. Dur-

ing soft-start, CSS is charged with a 6µA current. When the

LTC3805-5 comes out of shutdown, the LTC3805-5 quickly

charges CSS to about 0.7V at which point GATE begins

switching. From that point, GATE continues switching with

increasing duty cycle until the SSFLT pin reaches about

2.25V at which point soft-start is over and closed-loop

regulation begins. The voltage on the SSFLT pin addition-

ally further charges to about 4.75V.

CSS also performs the timeout function in the event of a

fault. After a fault, CSS is slowly discharged from about

4.75V to about 0.7V by a 2µA current. When the voltage

on the SSFLT pin reaches 0.7V the converter attempts to

restart. More detail on programming the external soft-start

fault timeout is described in the Applications Information

section.

Powering the LTC3805-5

A built-in shunt regulator from the VCC pin to GND limits

the voltage on the VCC pin to approximately 9.5V as long

as the shunt regulator is not forced to sink more than

25mA. The shunt regulator is always active, even when

the LTC3805-5 is in shutdown, since it serves the vital

function of protecting the VCC pin from overvoltage. The

shunt regulator permits the use of a wide variety of pow-

ering schemes for the LTC3805-5 even from high voltage

sources that exceed the LTC3805-5’s absolute maximum

ratings. Further details on powering schemes are described

in the Applications Information section.

Adjustable Slope Compensation

The LTC3805-5 injects a 10µA peak current ramp out of

its ISENSE pin which can be used, in conjunction with an

external resistor, for slope compensation in designs that

require it. This current ramp is approximately linear and

begins at zero current at 6% duty cycle, reaching peak

current at 80% duty cycle. Additional details are provided

in the Applications Information section.

LTC3805-5

38055fe

APPLICATIONS INFORMATION

Many LTC3805-5 application circuits can be derived from

the topologies shown on the ﬁrst page or in the Typical

Applications section of this data sheet.

The LTC3805-5 itself imposes no limits on allowed input

voltage VIN or output voltage VOUT. These are all determined

by the ratings of the external power components. In Figure 8,

the factors are: Q1 maximum drain-source voltage (BVDSS),

on-resistance (RDS(ON)) and maximum drain current, T1

saturation ﬂux level and winding insulation breakdown

voltages, CIN and COUT maximum working voltage, equiva-

lent series resistance (ESR), and maximum ripple current

ratings, and D1 and RSENSE power ratings.

VCC Bias Power

The VCC pin must be bypassed to the GND pin with a

minimum 1µF ceramic or tantalum capacitor located im-

mediately adjacent to the two pins. Proper supply bypassing

is necessary to supply the high transient currents required

by the MOSFET gate driver.

For maximum ﬂexibility, the LTC3805-5 is designed so

that it can be operated from voltages well beyond the

LTC3805-5’s absolute maximum ratings. Figure 2 shows

the simplest case, in which the LTC3805-5 is powered

with a resistor RVCC connected between the input voltage

and VCC. The built-in shunt regulator limits the voltage on

the VCC pin to around 9.5V as long as the internal shunt

regulator is not forced to sink more than 25mA. This pow-

ering scheme has the drawback that the power loss in the

resistor reduces converter efﬁciency and the 25mA shunt

regulator maximum may limit the maximum-to-minimum

range of input voltage.

The typical application circuit in Figure 9 shows a different

ﬂyback converter bias power strategy for a case in which

neither the input or output voltage is suitable for providing

bias power to the LTC3805-5. A small NPN preregulator

transistor and a Zener diode are used to accelerate the

rise of VCC and reduce the value of the VCC bias capacitor.

The ﬂyback transformer has an additional bias winding to

provide bias power. Note that this topology is very power-

ful because, by appropriate choice of transformer turns

ratio, the output voltage can be chosen without regard to

the value of the input voltage or the VCC bias power for

the LTC3805-5. The number of turns in the bias winding

is chosen according to

NBIAS =NSEC

VOUT +VD1

where NBIAS is the number of turns in the bias winding,

NSEC is the number of turns in the secondary winding,

VCC is the desired voltage to power the LTC3805-5, VOUT

is the converter output voltage, VD1 is the forward voltage

drop of D1 and VD2 is the forward voltage drop of D2.

Note that since VOUT is regulated by the converter control

loop, VCC is also regulated although not as precisely. If an

“off-the-shelf” transformer with excessive bias windings

is used, the resistor, RBIAS in Figure 9, can be added to

limit the current.

Transformer Design Considerations

Transformer speciﬁcation and design is perhaps the most

critical part of applying the LTC3805-5 successfully. In

addition to the usual list of caveats dealing with high fre-

quency power transformer design, the following should

prove useful.

Turns Ratios

Due to the use of the external feedback resistor divider

ratio to set output voltage, the user has relative freedom

in selecting transformer turns ratio to suit a given ap-

plication. Simple ratios of small integers, e.g., 1:1, 2:1,

3:2, etc. can be employed which yield more freedom in

setting total turns and transformer inductance. Simple

integer turns ratios also facilitate the use of “off-the-shelf”

conﬁgurable transformers. Turns ratio can be chosen on

Figure 2. Powering the LTC3805-5 via the

Internal Shunt Regulator

LTC3805-5

VCC

RVCC

CVCC

38055 F02

VIN

GND

LTC3805-5

38055fe

APPLICATIONS INFORMATION

the basis of desired duty cycle. However, remember that

the input supply voltage plus the secondary-to-primary

referred version of the ﬂyback pulse (including leakage

spike) must not exceed the allowed external MOSFET

breakdown rating.

Leakage Inductance

Transformer leakage inductance (on either the primary

or secondary) causes a voltage spike to occur after the

turn off of MOSFET (Q1) in Figure 8. This is increasingly

prominent at higher load currents, where more stored

energy must be dissipated. In some cases an RC “snubber”

circuit will be required to avoid overvoltage breakdown at

the MOSFET’s drain node. Application Note 19 is a good

reference on snubber design. A biﬁlar or similar winding

technique is a good way to minimize troublesome leak-

age inductances. However, remember that this will limit

the primary-to-secondary breakdown voltage, so biﬁlar

winding is not always practical.

Setting Undervoltage and Hysteresis on VIN

The RUN pin is connected to a resistive voltage divider

connected to VIN as shown in Figure 3. The voltage thresh-

old for the RUN pin is VRUNON rising and VRUNOFF falling.

Note that VRUNON – VRUNOFF = 35mV of built-in voltage

hysteresis that helps eliminate false trips.

To introduce further user-programmable hysteresis, the

LTC3805-5 sources 5µA out of the RUN pin when operation

of LTC3805-5 is enabled. As a result, the falling threshold

for the RUN pin also depends on the value of R1 and can

be programmed by the user. The falling threshold for VIN

is therefore

VIN(RUN,FALLING) =VRUNOFF •R1+R2

−R1•5µA

where R1(5µA) is the additional hysteresis introduced

by the 5µA current sourced by the RUN pin. When in

shutdown, the RUN pin does not source the 5µA current

and the rising threshold for VIN is simply

VIN(RUN,RISING) =VRUNON •

R1+R2

Note that for some applications the RUN pin can be con-

nected to VCC in which case the VCC thresholds, VTURNON

and VTURNOFF, control operation.

External Run/Stop Control

To implement external run control, place a small N-channel

MOSFET from the RUN pin to GND as shown in Figure 3.

Drive the gate of this MOSFET high to pull the RUN pin

to ground and prevent converter operation.

Selecting Feedback Resistor Divider Values

The regulated output voltage is determined by the resistor

divider across VOUT (R3 and R4 in Figure 8). The ratio

of R4 to R3 needed to produce a desired VOUT can be

calculated:

R3 =

OUT

−0.8V

0.8V R4

Choose resistance values for R3 and R4 to be as large as

possible in order to minimize any efﬁciency loss due to the

static current drawn from VOUT, but just small enough so

that when VOUT is in regulation the input current to the VFB

pin is less than 1% of the current through R3 and R4. A

good rule of thumb is to choose R4 to be less than 80k.

Figure 3. Setting RUN Pin Voltage and Run/Stop Control

LTC3805-5

RUN

RUN/STOP

CONTROL

(OPTIONAL)

GND 38055 F03

VIN

LTC3805-5

38055fe

APPLICATIONS INFORMATION

Feedback in Isolated Applications

Isolated applications do not use the FB pin and error ampli-

ﬁer but control the ITH pin directly using an opto-isolator

driven on the other side of the isolation barrier as shown

in Figure 4. For isolated converters, the FB pin is grounded

which provides pull-up on the ITH pin. This pull-up is not

enough to properly bias the opto-isolator which is typically

biased using a resistor to VCC. Since the ITH pin cannot

sink the opto-isolator bias current, a diode is required to

block it from the ITH pin. A low leakage Schottky diode,

or low forward voltage PN junction diode, should be used

to ensure that the opto-isolator is able to pull ITH down

to its lower clamp.

Oscillator Synchronization

The oscillator may be synchronized to an external clock

by connecting the synchronization signal to the SYNC pin.

The LTC3805-5 oscillator and turn-on of the switch are

synchronized to the rising edge of the external clock. The

frequency of the external sync signal must be ±33% with

respect to fOSC (as programmed by RFS). Additionally, the

value of fSYNC must be between 70kHz and 700kHz.

Current Sense Resistor Considerations

The external current sense resistor (RSENSE in Figure 8)

allows the user to optimize the current limit behavior for

the particular application. As the current sense resistor

is varied from several ohms down to tens of milliohms,

peak switch current goes from a fraction of an ampere to

several amperes. Care must be taken to ensure proper

circuit operation, especially with small current sense

resistor values.

For example, with the peak current sense voltage of 100mV

on the ISENSE pin, a peak switch current of 5A requires

a sense resistor of 0.020W. Note that the instantaneous

peak power in the sense resistor is 0.5W and it must be

rated accordingly. The LTC3805-5 has only a single sense

line to this resistor. Therefore, any parasitic resistance

in the ground side connection of the sense resistor will

increase its apparent value. In the case of a 0.020W sense

resistor, one milliohm of parasitic resistance will cause a

5% reduction in peak switch current. So the resistance of

printed circuit copper traces and vias cannot necessarily

be ignored.

Programmable Slope Compensation

The LTC3805-5 injects a ramping current through its ISENSE

pin into an external slope compensation resistor RSLOPE.

This current ramp starts at zero right after the GATE pin

has been high for the LTC3805-5’s minimum duty cycle

of 6%. The current rises linearly towards a peak of 10µA

at the maximum duty cycle of 80%, shutting off once the

GATE pin goes low. A series resistor RSLOPE connecting the

ISENSE pin to the current sense resistor RSENSE develops a

ramping voltage drop. From the perspective of the ISENSE

pin, this ramping voltage adds to the voltage across the

sense resistor, effectively reducing the current comparator

threshold in proportion to duty cycle. This stabilizes the

control loop against subharmonic oscillation. The amount

of reduction in the current comparator threshold (DVSENSE)

can be calculated using the following equation:

DVSENSE =DutyCycle −6%

80% 10µA •RSLOPE

Note: LTC3805-5 enforces 6% < Duty Cycle < 80%. A good

starting value for RSLOPE is 3k, which gives a 30mV drop

in current comparator threshold at 80% duty cycle.

Designs that do not operate at greater than 50% duty cycle

do not need slope compensation and may replace RSLOPE

with a direct connection.

Figure 4. Circuit for Isolated Feedback

LTC3805-5

ITH

VCC

ISOLATION

BARRIER

GND 38055 F04

LTC3805-5

38055fe

APPLICATIONS INFORMATION

Overcurrent Threshold Adjustment

Figure 5 shows the connection of the overcurrent pin, OC,

along with the ISENSE pin and the current sense resistor

RSENSE located in the source circuit of the power NMOS

which is driven by the GATE pin. The internal overcurrent

threshold on the OC pin is set at VOCT = 100 mV which is the

same as the peak current sense voltage VI(MAX) = 100 mV on

the ISENSE pin. The role of the slope compensation adjust-

ment resistor RSLOPE and the slope compensation current

ISLOPE is discussed in the prior section. In combination with

the overcurrent threshold adjust current IOC = 10µA, an

external resistor ROC can be used to lower the overcurrent

trip threshold from 100mV. This section describes how

to pick ROC to achieve the desired performance. In the

discussion that follows be careful to distinguish between

“current limit” where the converter continues to run with

the ISENSE pin limiting current on a cycle-by-cycle basis

while the output voltage falls below the regulation point

and “overcurrent protection” where the OC pin senses an

overcurrent and shuts down the converter for a timeout

period before attempting an automatic restart.

One overcurrent protection strategy is for the converter

to never enter current limit but to maintain output volt-

age regulation up to the point of tripping the overcurrent

protection. Operation at minimum input voltage VIN(MIN)

hits current limiting for the smallest output current and

is the design point for this strategy.

First, for operation at VIN(MIN), calculate the duty cycle Duty

Cycle VIN(MIN) using the appropriate formula depending on

whether the converter is a boost, ﬂyback or SEPIC. Then

use Duty Cycle VIN(MIN) to calculate DVSENSE(VIN(MIN))

using the formula in the prior section. For overcurrent

protection to trip at exactly the point where current limit-

ing would begin set:

ROC(CRIT) =

DVSENSE VIN(MIN)

( )

10µA

To ﬁnd the actual output current that trips overcurrent

protection, calculate the peak switch current IPK(VIN(MIN))

from:

IPK VIN(MIN)

( )

100mV − DVSENSE VIN(MIN)

( )

RSENSE

Then calculate the converter output current that corre-

sponds to IPK(VIN(MIN)). Again, the calculation depends

both on converter type and the details of converter design

including inductor current ripple. For minimum input volt-

age, ROC(CRIT) produces an overcurrent trip at an output

current just before loss of output voltage regulation and

the onset of current limiting. Note that the output current

that causes an overcurrent trip is higher for higher input

voltages but that an overcurrent trip will always occur

before loss of output voltage regulation. If desired to

meet a speciﬁc design target, an increase in ROC above

ROC(CRIT) can be used to reduce the trip threshold and

make the converter trip for a lower output current.

This calculation is based on steady-state operation. De-

pending on design, overcurrent protection can also be

triggered during a start-up transient, particularly if large

output ﬁlter capacitors are being charged as output voltage

rises. If that is a problem, output capacitor charging can

be slowed by using a larger value of SSFLT capacitor. It is

also possible to trip overcurrent protection during a load

step especially if the trip threshold is lowered by making

ROC > ROC(CRIT).

Another overcurrent protection strategy is keep the con-

verter running as current limiting reduces the duty cycle

and the output voltage sags. In this case, the goal is often

keep the converter in normal operation over as wide a range

as possible, including current limiting, and to trigger the

Figure 5. Circuit to Decrease Overcurrent Threshold

LTC3805-5

GATE

ISENSE

RSENSE

RSLOPE

IOC = 10µA

ISLOPE

GND 38055 F05

ROC

LTC3805-5

38055fe

APPLICATIONS INFORMATION

overcurrent trip only to prevent damage. To implement

this strategy use a value of ROC smaller than ROC(CRIT).

This also reduces sensitivity to overcurrent trips caused by

transient operation. In the limit, set ROC = 0 and connect

the OC pin directly to RSENSE. This causes an overcurrent

trip near minimum duty cycle or around 6%.

In some cases it may be desirable to increase the trip

threshold even further. In this strategy, the converter is

allowed to operate all the way down to minimum duty

cycle at which point the cycle-by-cycle current limit of

the ISENSE pin is lost and switch current goes up propor-

tionally to the output current. Figures 6 and 7 show two

ways to do this. Figure 6 is for relatively low currents with

relatively large values of RSENSE. Using this circuit the

overcurrent trip threshold is increased from 100mV to:

VOC =

SENSE1

SENSE2

RSENSE1

100mV

where it is assumed that the values of RSENSE1 and

RSENSE2 are so small that the IOC = 10µA threshold adjust-

ment current produces a negligible change in VOC.

For larger currents, values of the current sense resistors

must be very small and the circuit of Figure 6 becomes

impractical. The circuit of Figure 7 can be substituted and the

current sense threshold is increased from 100mV to:

VOC =R1

100mV

where the values of R1 and R2 should be kept below 10W

to prevent the IOC = 10µA threshold adjustment current

from producing a shift in VOC.

External Soft-Start Fault Timeout

The external soft-start is programmed by a capacitor CSS

from the SSFLT pin to GND. At the initiation of soft-start

the voltage on the SSFLT pin is quickly charged to 0.7V

at which point GATE begins switching. From that point,

a 6µA current charges the voltage on the SSFLT pin until

the voltage reaches about 2.25V at which point soft-start

is over and the converter enters closed-loop regulation.

The soft-start time tSS(EXT) as a function of the soft-start

capacitor CSS is therefore:

tSS(EXT) =CSS

2.25 −0.7V

6µA

After soft-start is complete, the voltage on the SSFLT pin

continues to charge to about a ﬁnal value of 4.75V. Note

that choosing a value of CSS less than 5.8nF has no effect

since it would attempt to program an external soft-start

time tSS(EXT) less than the mandatory minimum internal

soft-start time tSS(IN) = 1.8ms.

If there is an overcurrent fault detected on the OC pin, the

LTC3805-5 enters a shutdown mode while a 2µA current

discharges the voltage on the SSFLT pin from 4.75V to

about 0.7V. The fault timeout tFTO(EXT) is therefore:

tFTO(EXT) =CSS

4.75V −0.7V

2µA

At this point, the LTC3805-5 attempts a restart.

In the event of a persistent fault, such as a short-circuit

on the converter output, the converter enters a “hiccup”

mode where it continues to try and restart at repetition

rate determined by CSS. If the fault is eventually removed

the converter successfully restarts.

Figure 6. Circuit to Increase the Overcurrent

Threshold for Small Switch Currents

Figure 7. Circuit to Increase the Overcurrent

Threshold for Large Switch Currents

LTC3805-5

GATE

ISENSE

RSENSE1

RSENSE2

RSLOPE

GND 38055 F06

IOC = 10µA

ISLOPE LTC3805-5

GATE

ISENSE

RSLOPE

GND 38055 F07

RSENSE

IOC = 10µA

ISLOPE

LTC3805-5

38055fe

TYPICAL APPLICATIONS

Figure 8. 5.5V to 40V to 12V/2A SEPIC Converter

LTC3805-5

0.1µF 4.7µF

BAS516

10k

6.8nF

GND

ITH

SSFLT GATE

RUN

VCC

ISENSE

SYNC

10µF

50V

PDS760

4.56µH

BH510-1009

CIN

10µF

50V

COUT

47µF

16V

R4 7.15k

69.8k 80.6k

3.01k RSENSE

0.005Ω

38055 F08

HAT2266

VOUT

12V

100k

301W

221k

57.6k

VIN

5.5V T0 40V

PDZ6.8B

MMBT6428LT1

•

LOAD CURRENT (A)

0.01

EFFICIENCY (%)

POWER LOSS (W)

0.1 1 10

38055 TA04b

100

0.5

1.0

1.5

2.0

2.5

3.0

3.5

5.5V

10V

20V

30V

40V

EFFICIENCY

POWER LOSS

Efﬁciency and Power Loss vs Load Current

LTC3805-5

38055fe

TYPICAL APPLICATIONS

Figure 9. Isolated Telecom Supply: 18V to 72V Input to 3.3V/3A Output

Efﬁciency and Power Loss vs Load Current and VIN

LOAD CURRENT (A)

0.01

EFFICIENCY (%)

POWER LOSS (W)

100

0.1 1 10

38055 TA03b

2.0

3.0

4.0

1.0

1.5

2.5

3.5

0.5

18V

36V

48V

60V

72V

EFFICIENCY

POWER LOSS

VIN

COMP

GND

OPTO

75k15.8k

1µF

0.47µF

100k

47pF

22.1k

2200pF

250VAC

274Ω

2.2nF

221k

BAS516

FDC2512

VIN

VCC

PA1277NL

PDS1040

100µF

6.3V

100µF

6.3V

100µF

6.3V

RBIAS

402Ω

10W BAS516 BA760

68Ω

150pF

200V

PDZ6.8B

6.8V

MMBTA42

VIN–

6.8k

221k

LTC3805-5

ITH

GND

SSFLT

RUN

GATE

ISENSE

SYNCFS

VCC

56k

38055 F09

0.04Ω

VOUT+

3.3V

VOUT–

PS2801-1-K U2

LT4430

VIN+

18V TO

72V 2.2µF

100V

2.2µF

100V

0.022µF

1µF

221k

3.01k

LTC3805-5

38055fe

DD Package

10-Lead Plastic DFN (3mm × 3mm)

(Reference LTC DWG # 05-08-1699 Rev B)

MSE Package

10-Lead Plastic MSOP

(Reference LTC DWG # 05-08-1664 Rev C)

PACKAGE DESCRIPTION

3.00 ±0.10

(4 SIDES)

NOTE:

1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).

CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE

TOP AND BOTTOM OF PACKAGE

0.40 ± 0.10

BOTTOM VIEW—EXPOSED PAD

1.65 ± 0.10

(2 SIDES)

0.75 ±0.05

R = 0.125

TYP

2.38 ±0.10

(2 SIDES)

106

PIN 1

TOP MARK

(SEE NOTE 6)

0.200 REF

0.00 – 0.05

(DD) DFN REV B 0309

0.25 ± 0.05

2.38 ±0.05

(2 SIDES)

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

1.65 ±0.05

(2 SIDES)2.15 ±0.05

0.50

BSC

0.70 ±0.05

3.55 ±0.05

PACKAGE

OUTLINE

0.25 ± 0.05

0.50 BSC

MSOP (MSE) 0908 REV C

0.53 ± 0.152

(.021 ± .006)

SEATING

PLANE

0.18

(.007)

1.10

(.043)

MAX

0.17 – 0.27

(.007 – .011)

TYP

0.86

(.034)

REF

0.50

(.0197)

BSC

1234 5

4.90 ± 0.152

(.193 ± .006)

0.497 ± 0.076

(.0196 ± .003)

REF

8910

3.00 ± 0.102

(.118 ± .004)

(NOTE 3)

3.00 ± 0.102

(.118 ± .004)

(NOTE 4)

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

0.254

(.010) 0° – 6° TYP

DETAIL “A”

GAUGE PLANE

5.23

(.206)

MIN

3.20 – 3.45

(.126 – .136)

0.889 ± 0.127

(.035 ± .005)

RECOMMENDED SOLDER PAD LAYOUT

0.305 ± 0.038

(.0120 ± .0015)

TYP

2.083 ± 0.102

(.082 ± .004)

2.794 ± 0.102

(.110 ± .004)

0.50

(.0197)

BSC

BOTTOM VIEW OF

EXPOSED PAD OPTION

1.83 ± 0.102

(.072 ± .004)

2.06 ± 0.102

(.081 ± .004)

0.1016 ± 0.0508

(.004 ± .002)

DETAIL “B”

CORNER TAIL IS PART OF

THE LEADFRAME FEATURE.

FOR REFERENCE ONLY

NO MEASUREMENT PURPOSE

0.05 REF

0.29

REF

LTC3805-5

38055fe

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER

D 04/10 Updates to Absolute Maximum Ratings

Changes to Electrical Characteristics Table

Updates to Note 2

Update to Pin 11 in Pin Functions

Edits to Start-Up Shutdown in Operation Section

Change to Typical Application

Updated Related Parts Table

17, 20

E 05/11 Added MP-grade part. Reﬂected throughout the data sheet 1-20

(Revision history begins at Rev D)

LTC3805-5

38055fe

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

 LINEAR TECHNOLOGY CORPORATION 2008

LT 0511 REV E • PRINTED IN USA

RELATED PARTS

TYPICAL APPLICATION

5V to 40V to 12V/1A Nonisolated Flyback Converter

LOAD CURRENT (A)

0.01

EFFICIENCY (%)

POWER LOSS (W)

0.1 1 10

38055 TA02b

5.5V

10V

20V

30V

40V

Efﬁciency and Power Loss

vs Load Current

75k

3.01k

4.7µF

100µF

16V

51.1k

47pF

CSS

0.1µF

UPS840

FDC2512

VCC

220Ω

150pF

200V

38055 TA02

2.2µF

100V

VIN

5V TO 40V

VCC

LTC3805-5

ITH

GND

SSFLT

RUN

GATE

ISENSE

SYNCFS

VCC

100µF

16V

RSENSE

0.005Ω

VOUT

12V

470pF

100pF

20k 3.65k

2.2µF

100V 10k

6.8V

PDZ6.8B 680Ω

BAS516

T3772

MMBTA42

PART NUMBER DESCRIPTION COMMENTS

LT3748 100V No Opto Flyback Controller 5V ≤ VIN ≤ 100V, Boundary Mode Operation, MSOP-16 with Extra High

Voltage Pin Spacing

LT3758 Boost, Flyback, SEPIC and Inverting Controller 5.5V ≤ VIN ≤ 100V, 100kHz to 1MHz Programmable Operating Frequency,

3mm × 3mm DFN-10 and MSOP-10E

LT3575 No Opto-Isolated Flyback with 60V Integrated Switch 3V ≤ VIN ≤ 40V, Up to 14W, Boundary Mode Operation, TSSOP-16E

LTC3803/LTC3803-3/

LTC3803-5

Flyback DC/DC Controller with Fixed 200kHz or

300kHz Operating Frequency

VIN and VOUT Limited Only by External Components, 6-Pin ThinSOT™

Package

LTC3873/LTC3873-5 No RSENSE™ Constant Frequency Flyback, Boost,

SEPIC Controller

VIN and VOUT Limited Only by External Components, 8-Pin ThinSOT

or 2mm × 3mm DFN-8 Packages

LT3825 No Opto-Isolated Synchronous Flyback Controller VIN 16V to 75V Limited by External Components, Up to 60W, TSSOP-16E