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nDIM
LM3429
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PGND
NC
DAP
GATE
COMP
VIN
CSH
RCT
IS
HSP
VCC
HSN
1
2
3
4
5
6
7
14
13
12
11
10
9
8
ILED
PWM
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LM3429/-Q1 N-Channel Controller for Constant Current LED Drivers
1 Features 3 Description
The LM3429 is a versatile high voltage N-channel
1• LM3429-Q1 is AEC-Q100 Grade 1 Qualified for MosFET controller for LED drivers. It can be easily
Automotive Applications configured in buck, boost, buck-boost and SEPIC
• VIN Range From 4.5 V to 75 V topologies. This flexibility, along with an input voltage
• Adjustable Current Sense Voltage rating of 75V, makes the LM3429 ideal for
illuminating LEDs in a very diverse, large family of
• High-Side Current Sensing applications.
• 2-Ω, 1-A Peak MosFET Gate Driver Adjustable high-side current sense voltage allows for
• Input Undervoltage Protection tight regulation of the LED current with the highest
• Overvoltage Protection efficiency possible. The LM3429 uses Predictive Off-
• PWM Dimming time (PRO) control, which is a combination of peak
current-mode control and a predictive off-timer. This
• Analog Dimming method of control eases the design of loop
• Cycle-by-Cycle Current Limit compensation while providing inherent input voltage
• Programmable Switching Frequency feed-forward compensation.
• Low Profile 14-lead HTSSOP Package The LM3429 includes a high-voltage startup regulator
• Thermal Shutdown that operates over a wide input range of 4.5 V to 75
V. The internal PWM controller is designed for
2 Applications adjustable switching frequencies of up to 2 MHz, thus
enabling compact solutions. Additional features
• LED Drivers - Buck, Boost, Buck-Boost, SEPIC include analog dimming, PWM dimming, overvoltage
• Indoor and Outdoor SSL protection, undervoltage lock-out, cycle-by-cycle
current limit, and thermal shutdown.
• Automotive
• General Illumination Device Information(1)
• Constant-Current Regulators PART NUMBER PACKAGE BODY SIZE (NOM)
LM3429 HTSSOP (14) 5.00 mm × 4.40 mm
LM3429-Q1
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Boost Application Circuit
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
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Table of Contents
1 Features.................................................................. 18 Application and Implementation ........................ 22
8.1 Application Information............................................ 22
2 Applications ........................................................... 18.2 Typical Applications ................................................ 24
3 Description............................................................. 19 Power Supply Recommendations...................... 53
4 Revision History..................................................... 29.1 Input Supply Current Limit ...................................... 53
5 Pin Configuration and Functions......................... 310 Layout................................................................... 53
6 Specifications......................................................... 410.1 Layout Guidelines ................................................. 53
6.1 Absolute Maximum Ratings ...................................... 410.2 Layout Example .................................................... 54
6.2 ESD Ratings.............................................................. 411 Device and Documentation Support................. 55
6.3 Recommended Operating Conditions....................... 411.1 Device Support...................................................... 55
6.4 Thermal Information.................................................. 511.2 Documentation Support ........................................ 55
6.5 Electrical Characteristics .......................................... 511.3 Related Links ........................................................ 55
6.6 Typical Characteristics ............................................. 711.4 Community Resources.......................................... 55
7 Detailed Description.............................................. 911.5 Trademarks........................................................... 55
7.1 Overview................................................................... 911.6 Electrostatic Discharge Caution............................ 56
7.2 Functional Block Diagram......................................... 911.7 Glossary................................................................ 56
7.3 Feature Description................................................. 10 12 Mechanical, Packaging, and Orderable
7.4 Device Functional Modes........................................ 21 Information ........................................................... 56
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (April 2013) to Revision H Page
• Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes,Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
Changes from Revision F (May 2013) to Revision G Page
• Changed layout of National Data Sheet to TI format ........................................................................................................... 51
• Changed layout of National Data Sheet to TI format ........................................................................................................... 52
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VIN
nDIM
AGND
OVP
CSH
RCT
GATE
1
COMP HSP
DAP
IS
2
3
4
5
6
7
14
13
12
11
10
9
8NC
PGND
HSN
VCC
15
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SNVS616H –APRIL 2009–REVISED JULY 2015
5 Pin Configuration and Functions
PWP Package
14- Pin HTSSOP
Top View
Pin Functions
PIN I/O DESCRIPTION APPLICATION INFORMATION
NO. NAME
Bypass with 100 nF capacitor to AGND as close to the device as
1 VIN I Input Voltage possible in the circuit board layout.
2 COMP I Compensation Connect a capacitor to AGND to set compensation.
Connect a resistor to AGND to set signal current. For analog
3 CSH I Current Sense High dimming, connect current source or potentiometer to AGND (see
Analog Dimming section).
Connect a resistor from the switch node and a capacitor to AGND to
4 RCT I Resistor Capacitor Timing set the switching frequency.
Connect to PGND through the DAP copper circuit board pad to
5 AGND GND Analog Ground provide proper ground return for CSH, COMP, and RCT.
Connect to a resistor divider from the output (VO) or the input to
program output overvoltage lockout (OVLO). Turn-off threshold is
6 OVP I Overvoltage Protection 1.24 V and hysteresis for turn-on is provided by 20 µA current
source.
Connect a PWM signal for dimming as detailed in the PWM Dimming
section and/or a resistor divider from VIN to program input
7 nDIM I Not DIM input undervoltage lockout (UVLO). Turn-on threshold is 1.24 V and
hysteresis for turn-off is provided by 20 µA current source.
8 NC No Connection Leave open.
Connect to AGND through DAP copper pad to provide ground return
9 PGND GND Power Ground for GATE.
10 GATE O Gate Drive Output Connect to the gate of the external NFET.
11 VCC I Internal Regulator Output Bypass with a 2.2 µF–3.3 µF, ceramic capacitor to PGND.
Connect to the drain of the main N-channel MosFET switch for RDS-
12 IS I Main Switch Current Sense ON sensing or to a sense resistor installed in the source of the same
device.
Connect through a series resistor to LED current sense resistor
13 HSP I LED Current Sense Positive (positive).
Connect through a series resistor to LED current sense resistor
14 HSN I LED Current Sense Negative (negative).
DAP DAP GND Thermal pad on bottom of IC Connect to AGND and PGND.
(15)
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)(2)
MIN MAX UNIT
VIN, nDIM –0.3 76
OVP, HSP, HSN –0.3 76
RCT –0.3 3
–0.3 76
IS –2 for 100 ns V
Voltage VCC –0.3 8
COMP, CSH –0.3 6
–0.3 VCC
GATE –2.5 for 100 ns VCC+2.5 for 100 ns
–0.3 0.3
PGND –2.5 2.5 for 100 ns
VIN, nDIM –1 mA
OVP, HSP, HSN –100 µA
RCT –1 5
Continuous Current mA
IS –1
COMP, CSH –200 200 µA
GATE –1 1 mA
Maximum Junction Temperature Internally Limited
Maximum Lead Temperature (Reflow and Solder) (3) 260 °C
Continuous Power Dissipation Internally Limited
Storage Temperature, Tstg –65 150 °C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Refer to http://www.ti.com/packaging for more detailed information and mounting techniques.
6.2 ESD Ratings VALUE UNIT
LM3429 IN PWP PACKAGE
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins(1) ±2000
V(ESD) Electrostatic discharge V
Charged device model (CDM), per JEDEC specification JESD22-C101, all ±1000
pins(2)
LM3429-Q1 IN PWP PACKAGE
Human body model (HBM), per AEC Q100-002(3) ±2000
V(ESD) Electrostatic discharge V
Charged device model (CDM), per AEC Q100-011 ±1000
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
(3) AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions MIN MAX UNIT
Operating Junction Temperature Range –40 125 °C
Input Voltage VIN 4.5 75 V
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6.4 Thermal Information LM3429-Q1 LM3429
THERMAL METRIC(1) PWP (HTSSOP) PWP (HTSSOP) UNIT
14 PINS 14 PINS
RθJA Junction-to-ambient thermal resistance 47.8 47.8 °C/W
RθJC(top) Junction-to-case (top) thermal resistance 26.5 26.5 °C/W
RθJB Junction-to-board thermal resistance 22.3 22.3 °C/W
ψJT Junction-to-top characterization parameter 0.7 0.7 °C/W
ψJB Junction-to-board characterization parameter 22.1 22.1 °C/W
RθJC(bot) Junction-to-case (bottom) thermal resistance 3.3 3.3 °C/W
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
6.5 Electrical Characteristics
MIN and MAX limits apply TJ= (−40°C to 125°C) unless specified otherwise. Unless otherwise stated the following condition
applies: VIN = 14 V.
PARAMETER TEST CONDITIONS MIN(1) TYP(2) MAX(1) UNIT
STARTUP REGULATOR (VCC)
VCC-REG VCC Regulation ICC = 0 mA 6.3 6.9 7.35 V
ICC-LIM VCC Current Limit VCC = 0V 20 27 mA
IQQuiescent Current Static 1.6 3
VCC-UVLO VCC UVLO Threshold VCC Increasing 4.17 4.5
VCC Decreasing 3.7 4.08 V
VCC-HYS VCC UVLO Hysteresis 0.1
OVERVOLTAGE PROTECTION (OVP)
VTH-OVP OVP OVLO Threshold OVP Increasing 1.18 1.24 1.28 V
IHYS-OVP OVP Hysteresis Source Current OVP Active (high) 10 20 30 µA
ERROR AMPLIFIER
VCSH CSH Reference Voltage With Respect to AGND 1.21 1.235 1.26 V
Error Amplifier Input Bias Current MIN, MAX: TJ= 25°C –0.6 0 0.6 µA
COMP Sink / Source Current 10 26 40
Transconductance 100 µA/V
Linear Input Range (3) ±125 mV
Transconductance Bandwidth -6dB Unloaded
Response(3), MIN: TJ= 0.5 1 MHz
25°C
OFF TIMER (RCT)
tOFF-MIN Minimum Off-time RCT = 1V through 1 kΩ35 75 ns
RRCT RCT Reset Pulldown Resistance 36 120 Ω
VRCT VIN/25 Reference Voltage VIN = 14V 540 565 585 mV
PWM COMPARATOR
COMP to PWM Offset 700 800 900 mV
CURRENT LIMIT (IS)
VLIM Current Limit Threshold 215 245 275 mV
VLIM Delay to Output 35 75 ns
tON-MIN Leading Edge Blanking Time 75 250 450
(1) All limits specified at room temperature (TYP) and at temperature extremes (MIN/MAX). All room temperature limits are 100%
production tested. All limits at temperature extremes are specified through correlation using standard Statistical Quality Control (SQC)
methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) These electrical parameters are specified by design, and are not verified by test.
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Electrical Characteristics (continued)
MIN and MAX limits apply TJ= (−40°C to 125°C) unless specified otherwise. Unless otherwise stated the following condition
applies: VIN = 14 V.
PARAMETER TEST CONDITIONS MIN(1) TYP(2) MAX(1) UNIT
HIGH SIDE TRANSCONDUCTANCE AMPLIFIER
Input Bias Current 10 µA
Transconductance 20 119 mA/V
Input Offset Current –1.5 0 1.5 µA
Input Offset Voltage –7 0 7 mV
Transconductance Bandwidth ICSH = 100 µA(3), MIN: TJ250 500 kHz
= 25°C
GATE DRIVER (GATE)
RSRC(GATE) GATE Sourcing Resistance GATE = High 2 6 Ω
RSNK(GATE) GATE Sinking Resistance GATE = Low 1.3 4.5
UNDERVOLTAGE LOCKOUT and DIM INPUT (nDIM)
VTH-nDIM nDIM / UVLO Threshold 1.18 1.24 1.28 V
IHYS-nDIM nDIM Hysteresis Current 10 20 30 µA
THERMAL SHUTDOWN
TSD Thermal Shutdown Threshold (3) 165 °C
THYS Thermal Shutdown Hysteresis (3) 25
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DUTY CYCLE (%)
ILED (A)
1.0
0.8
0.6
0.4
0.2
0.00 20 40 60 80 100
100 Hz
500 Hz
ICSH (éA)
ILED (A)
1.0
0.8
0.6
0.4
0.2
0.0 0 20 40 60 80 100
VIN (V)
ILED (A)
1.00
0.99
0.98
0.97
0.96 5 10 15 20 25 30
VIN (V)
ILED (A)
1.05
1.03
1.01
0.99
0.97 0 16 32 48 64 80
VIN (V)
EFFICIENCY (%)
100
95
90
85
80 10 15 20 25 30
VIN (V)
EFFICIENCY (%)
100
95
90
85
80
75
70 0 16 32 48 64 80
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6.6 Typical Characteristics
TA= 25°C and VIN = 14 V unless otherwise specified. The measurements for Figure 1 and Figure 3 were made using the
standard boost evaluation board from AN-1986 (SNVA404). The measurements for Figure 2,Figure 4, and Figure 5,Figure 6
were made using the standard buck-boost evaluation board from AN-1985 (SNVA403).
Figure 2. Buck-Boost Efficiency vs Input Voltage
Figure 1. Boost Efficiency vs Input Voltage VO= 20 V (6 LEDs)
VO= 32 V (9 LEDs)
Figure 3. Boost LED Current vs Input Voltage Figure 4. Buck-boost LED Current vs Input Voltage
VO= 32 V (9 LEDs) VO= 20 V (6 LEDs)
Figure 5. Analog Dimming Figure 6. PWM Dimming
VO= 20 V (6 LEDs) VO= 20V (6 LEDs)
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TEMPERATURE (°C)
tON-MIN (ns)
280
275
270
265
260
255
250-50 -14 22 58 94 130
TEMPERATURE (°C)
VRCT (mV)
569
568
567
566
565
564-50 -14 22 58 94 130
TEMPERATURE (°C)
VLIM (mV)
246
244
242
240
238-50 -14 22 58 94 130
TEMPERATURE (°C)
VCSH (V)
1.250
1.245
1.240
1.235
1.230
1.225
1.220-50 -14 22 58 94 130
TEMPERATURE (°C)
VCC (V)
7.10
7.05
7.00
6.95
6.90
6.85
6.80-50 -14 22 58 94 130
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SNVS616H –APRIL 2009–REVISED JULY 2015
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Typical Characteristics (continued)
TA= 25°C and VIN = 14 V unless otherwise specified. The measurements for Figure 1 and Figure 3 were made using the
standard boost evaluation board from AN-1986 (SNVA404). The measurements for Figure 2,Figure 4, and Figure 5,Figure 6
were made using the standard buck-boost evaluation board from AN-1985 (SNVA403).
Figure 7. VCSH vs. Junction Temperature Figure 8. VCC vs. Junction Temperature
Figure 10. VLIM vs. Junction Temperature
Figure 9. VRCT vs. Junction Temperature
Figure 11. tON-MIN vs. Junction Temperature
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IS
PWM
Start new on time
nDIM
UVLO
HYSTERESIS
COMP
REFERENCE
CURRENT
LIMIT
CSH
HSN
LOGIC
HSP
S
R
Q
QB
UVLO
GATE
PGND
VccUVLO
Thermal
Limit
TLIM
VIN/25
Standby
Dimming
Reset
Dominant
RCT
LEB
AGND
VCC
VCC
VIN
6.9V LDO
Regulator
OVP
OVP
HYSTERESIS
20 PA
1.24V
800 mV
1.24V
20 PA
1.24V
1.24V
245 mV
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7 Detailed Description
7.1 Overview
The LM3429 is an N-channel MosFET (NFET) controller for buck, boost and buck-boost current regulators which
are ideal for driving LED loads. The controller has wide input voltage range allowing for regulation of a variety of
LED loads. The high-side differential current sense, with low adjustable threshold voltage, provides an excellent
method for regulating output current while maintaining high system efficiency. The LM3429 uses a Predictive Off-
time (PRO) control architecture that allows the regulator to be operated using minimal external control loop
compensation, while providing an inherent cycle-by-cycle current limit. The adjustable current sense threshold
provides the capability to amplitude (analog) dim the LED current and the output enable/disable function allows
for PWM dimming using no external components. When designing, the maximum attainable LED current is not
internally limited because the LM3429 is a controller. Instead it is a function of the system operating point,
component choices, and switching frequency allowing the LM3429 to easily provide constant currents up to 5A.
This simple controller contains all the features necessary to implement a high-efficiency versatile LED driver.
7.2 Functional Block Diagram
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D=INO VV +
O
V
D= INO VV -
O
V
D= O
V
IN
V
t
iL (t)
ÂiL-PP
IL-MAX
IL-MIN
IL
0
TS
tON = DTStOFF = (1-D)TS
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SNVS616H –APRIL 2009–REVISED JULY 2015
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7.3 Feature Description
7.3.1 Current Regulators
Current regulators can be designed to accomplish three basic functions: buck, boost, and buck-boost. All three
topologies in their most basic form contain a main switching MosFET, a recirculating diode, an inductor and
capacitors. The LM3429 is designed to drive a ground referenced NFET which is perfect for a standard boost
regulator. Buck and buck-boost regulators, on the other hand, usually have a high-side switch. When driving an
LED load, a ground referenced load is often not necessary, therefore a ground referenced switch can be used to
drive a floating load instead. The LM3429 can then be used to drive all three basic topologies as shown in the
Typical Applications section.
Looking at the buck-boost design, the basic operation of a current regulator can be analyzed. During the time
that the NFET (Q1) is turned on (tON), the input voltage source stores energy in the inductor (L1) while the output
capacitor (CO) provides energy to the LED load. When Q1 is turned off (tOFF), the re-circulating diode (D1)
becomes forward biased and L1 provides energy to both COand the LED load. Figure 12 shows the inductor
current (iL(t)) waveform for a regulator operating in CCM.
Figure 12. Ideal CCM Regulator Inductor Current iL(t)
The average output LED current (ILED) is proportional to the average inductor current (IL) , therefore if ILis tightly
controlled, ILED will be well regulated. As the system changes input voltage or output voltage, the ideal duty cycle
(D) is varied to regulate ILand ultimately ILED. For any current regulator, D is a function of the conversion ratio:
Buck
(1)
Boost
(2)
Buck-Boost
(3)
7.3.2 Predictive Off-Time (PRO) Control
PRO control is used by the LM3429 to control ILED. It is a combination of average peak current control and a one-
shot off-timer that varies with input voltage. The LM3429 uses peak current control to regulate the average LED
current through an array of HBLEDs. This method of control uses a series resistor in the LED path to sense LED
current and can use either a series resistor in the MosFET path or the MosFET RDS-ON for both cycle-by-cycle
current limit and input voltage feed forward. D is indirectly controlled by changes in both tOFF and tON, which vary
depending on the operating point.
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RT
CT
VSW
LM3429
RCT
Start tON
VIN/25
Reset timer
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Feature Description (continued)
Even though the off-time control is quasi-hysteretic, the input voltage proportionality in the off-timer creates an
essentially constant switching frequency over the entire operating range for boost and buck-boost topologies.
The buck topology can be designed to give constant ripple over either input voltage or output voltage, however
switching frequency is only constant at a specific operating point .
This type of control minimizes the control loop compensation necessary in many switching regulators, simplifying
the design process. The averaging mechanism in the peak detection control loop provides extremely accurate
LED current regulation over the entire operating range.
PRO control was designed to mitigate “current mode instability” (also called “sub-harmonic oscillation”) found in
standard peak current mode control when operating near or above 50% duty cycles. When using standard peak
current mode control with a fixed switching frequency, this condition is present, regardless of the topology.
However, using a constant off-time approach, current mode instability cannot occur, enabling easier design and
control.
Predictive off-time advantages:
• There is no current mode instability at any duty cycle.
• Higher duty cycles / voltage transformation ratios are possible, especially in the boost regulator.
The only disadvantage is that synchronization to an external reference frequency is generally not available.
7.3.3 Switching Frequency
An external resistor (RT) connected between the RCT pin and the switch node (where D1, Q1, and L1 connect),
in combination with a capacitor (CT) between the RCT and AGND pins, sets the off-time (tOFF) as shown in
Figure 13. For boost and buck-boost topologies, the VIN proportionality ensures a virtually constant switching
frequency (fSW).
Figure 13. Off-timer Circuitry for Boost and Buck-boost Regulators
For a buck topology, RTand CTare also used to set tOFF, however the VIN proportionality will not ensure a
constant switching frequency. Instead, constant ripple operation can be achieved. Changing the connection of RT
in Figure 13 from VSW to VIN will provide a constant ripple over varying VIN. Adding a PNP transistor as shown in
Figure 14 will provide constant ripple over varying VO.
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VSNS = 1.24V x RCSH
RHSP
ICSH = RHSP
VSNS
TT
SW CR 25
fx
=
( )
2
OOIN VVV25 -
xx
SW
f=2
INTT VCR xx
fSW = 25 x VIN - VO
RT x CT X VIN
( )
RT
CT
LM3429
RCT
Start tON
VIN/25
Reset timer
RSNS
VIN
LED-
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Feature Description (continued)
Figure 14. Off-timer Circuitry for Buck Regulators
The switching frequency is defined:
Buck (Constant Ripple vs. VIN)
(4)
Buck (Constant Ripple vs. VO)
(5)
Boost and Buck-Boost
(6)
For all topologies, the CTcapacitor is recommended to be 1 nF and should be located very close to the LM3429.
7.3.4 Average LED Current
The LM3429 uses an external current sense resistor (RSNS) placed in series with the LED load to convert the
LED current (ILED) into a voltage (VSNS) as shown in Figure 15. The HSP and HSN pins are the inputs to the
high-side sense amplifier which are forced to be equal potential (VHSP=VHSN) through negative feedback.
Because of this, the VSNS voltage is forced across RHSP to generate the signal current (ICSH) which flows out of
the CSH pin and through the RCSH resistor. The error amplifier will regulate the CSH pin to 1.24 V, therefore ICSH
can be calculated:
(7)
This means VSNS will be regulated as follows:
(8)
ILED can then be calculated:
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RSNS
ILED RHSP
RHSN HSN
HSP High-Side
Sense Amplifier
CSH 1.24V
CCMP
RCSH
COMP
Error Amplifier
VSNS
To PWM
Comparator
LM3429
ICSH
ILED = RSNS
1.24V
RSNS
VSNS RCSH
RHSP
= x
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Feature Description (continued)
(9)
The selection of the three resistors (RSNS, RCSH, and RHSP) is not arbitrary. For matching and noise performance,
the suggested signal current ICSH is approximately 100 µA. This current does not flow in the LEDs and will not
affect either the off state LED current or the regulated LED current. ICSH can be above or below this value, but
the high-side amplifier offset characteristics may be affected slightly. In addition, to minimize the effect of the
high-side amplifier voltage offset on LED current accuracy, the minimum VSNS is suggested to be 50 mV. Finally,
a resistor (RHSN = RHSP) should be placed in series with the HSN pin to cancel out the effects of the input bias
current (~10 µA) of both inputs of the high-side sense amplifier. The CSH pin can also be used as a low-side
current sense input regulated to 1.24 V. The high-side sense amplifier is disabled if HSP and HSN are tied to
GND.
Figure 15. LED Current Sense Circuitry
7.3.5 Analog Dimming
The CSH pin can be used to analog dim the LED current by adjusting the current sense voltage (VSNS). There
are several different methods to adjust VSNS using the CSH pin:
1. External variable resistance : Adjust a potentiometer placed in series with RCSH to vary VSNS.
2. External variable current source: Source current (0 µA to ICSH) into the CSH pin to adjust VSNS.
In general, analog dimming applications require a lower switching frequency to minimize the effect of the leading
edge blanking circuit. As the LED current is reduced, the output voltage and the duty cycle decreases.
Eventually, the minimum on-time is reached. The lower the switching frequency, the wider the linear dimming
range. Figure 16 shows how both methods are physically implemented.
Method 1 uses an external potentiometer in the CSH path which is a simple addition to the existing circuitry.
However, the LEDs cannot dim completely because there is always some resistance causing signal current to
flow. This method is also susceptible to noise coupling at the CSH pin because the potentiometer increases the
size of the signal current loop.
Method 2 provides a complete dimming range and better noise performance, though it is more complex. It
consists of a PNP current mirror and a bias network consisting of an NPN, 2 resistors and a potentiometer
(RADJ), where RADJ controls the amount of current sourced into the CSH pin. A higher resistance value will source
more current into the CSH pin causing less regulated signal current through RHSP, effectively dimming the LEDs.
VREF should be a precise external voltage reference, while Q7 and Q8 should be a dual pair PNP for best
matching and performance. The additional current (IADD) sourced into the CSH pin can be calculated:
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LM3429 LM3429-Q1
CSH
RCSH
LM3429
VCC
RBIAS
RMAX
Q6
Q7
VREF
RADJ
Q8
RADJ
Variable Current Source
Variable
Resistance
RHSP
ILED = ICSH - IADD x RSNS ¸
¹
·
¨
©
§
( )
RADJ x VREF
IADD = RADJ + RMAX - VBE-Q6
¸
¹
·
¨
©
§
RBIAS
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Feature Description (continued)
(10)
The corresponding ILED for a specific IADD is:
(11)
Figure 16. Analog Dimming Circuitry
7.3.6 Current Sense and Current Limit
The LM3429 achieves peak current mode control using a comparator that monitors the MosFET transistor
current, comparing it with the COMP pin voltage as shown in Figure 17. Further, it incorporates a cycle-by-cycle
overcurrent protection function. Current limit is accomplished by a redundant internal current sense comparator.
If the voltage at the current sense comparator input (IS) exceeds 245 mV (typical), the on cycle is immediately
terminated. The IS input pin has an internal N-channel MosFET which pulls it down at the conclusion of every
cycle. The discharge device remains on an additional 250 ns (typical) after the beginning of a new cycle to blank
the leading edge spike on the current sense signal. The leading edge blanking (LEB) determines the minimum
achievable on-time (tON-MIN).
14 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
x
=
¨
¨
©
§+s
1Z1P ¸
¸
¹
·
0U
T
U
T¨
¨
©
§-s
1Z1Z ¸
¸
¹
·
ILIM = 245 mV
RLIM
LM3429
IT
PWM
COMP
IS
RLIM
Q1 GATE
LEB
PGND
245 mV
800 mV
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Feature Description (continued)
Figure 17. Current Sense / Current Limit Circuitry
There are two possible methods to sense the transistor current. The RDS-ON of the main power MosFET can be
used as the current sense resistance because the IS pin was designed to withstand the high voltages present on
the drain when the MosFET is in the off state. Alternatively, a sense resistor located in the source of the MosFET
may be used for current sensing, however a low inductance (ESL) type is suggested. The cycle-by-cycle current
limit (ILIM) can be calulated using either method as the limiting resistance (RLIM):
(12)
In general, the external series resistor allows for more design flexibility, however it is important to ensure all of
the noise sensitive low power ground connections are connected together local to the controller and a single
connection is made to the high current PGND (sense resistor ground point).
7.3.7 Control Loop Compensation
The LM3429 control loop is modeled like any current mode controller. Using a first order approximation, the
uncompensated loop can be modeled as a single pole created by the output capacitor and, in the boost and
buck-boost topologies, a right half plane zero created by the inductor, where both have a dependence on the
LED string dynamic resistance. There is also a high frequency pole in the model, however it is above the
switching frequency and plays no part in the compensation design process therefore it will be neglected.
Because ceramic capacitance is recommended for use with LED drivers due to long lifetimes and high ripple
current rating, the ESR of the output capacitor can also be neglected in the loop analysis. Finally, there is a DC
gain of the uncompensated loop which is dependent on internal controller gains and the external sensing
network.
A buck-boost regulator will be used as an example case. See the Typical Applications section for compensation
of all topologies.
The uncompensated loop gain for a buck-boost regulator is given by the following equation:
(13)
Where the uncompensated DC loop gain of the system is described as:
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LM3429 LM3429-Q1
PHASE (°)
FREQUENCY (Hz)
GAIN (dB)
100
80
60
40
20
0
-20
-40
-60
135
90
45
0
-45
-90
-135
-180
-225
1e-1 1e1 1e3 1e5 1e7
0° Phase Margin
öP1
PHASE
GAIN
öZ1
=Dr 2
Dc
x
1Z
ZL1Dx
1P =Z1+D
OD Cr x
3
=
0U
T=SNSCSH RR500VD xxx
c620VD x
c
( ) LIM
LED RID1 xx+
( ) LIMHSP RRD1 xx+
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Feature Description (continued)
(14)
And the output pole (ωP1) is approximated:
(15)
And the right half plane zero (ωZ1) is:
(16)
Figure 18. Uncompensated Loop Gain Frequency Response
Figure 18 shows the uncompensated loop gain in a worst-case scenario when the RHP zero is below the output
pole. This occurs at high duty cycles when the regulator is trying to boost the output voltage significantly. The
RHP zero adds 20dB/decade of gain while loosing 45°/decade of phase which places the crossover frequency
(when the gain is zero dB) extremely high because the gain only starts falling again due to the high frequency
pole (not modeled or shown in figure). The phase will be below -180° at the crossover frequency which means
there is no phase margin (180° + phase at crossover frequency) causing system instability. Even if the output
pole is below the RHP zero, the phase will still reach -180° before the crossover frequency in most cases yielding
instability.
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Product Folder Links: LM3429 LM3429-Q1
x= 0U
TT -1 ¸
¸
¹
·
¨
¨
©
§s
Z1
Z
xx +1 ¸
¸
¹
·
¨
¨
©
§s
Z3P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z2P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z1P
1
3P =Z
FSFS CR x
P2 6CMP
1
5x10 x C
Z
:
RSNS
ILED RHSP
RHSN HSN
HSP High-Side
Sense Amplifier
CSH 1.24V
CCMP
RCSH
COMP
Error Amplifier
VSNS
To PWM
Comparator
LM3429
CFS
RFS
sets öP3
RO
sets öP2
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Feature Description (continued)
Figure 19. Compensation Circuitry
To mitigate this problem, a compensator should be designed to give adequate phase margin (above 45°) at the
crossover frequency. A simple compensator using a single capacitor at the COMP pin (CCMP) will add a dominant
pole to the system, which will ensure adequate phase margin if placed low enough. At high duty cycles (as
shown in Figure 18), the RHP zero places extreme limits on the achievable bandwidth with this type of
compensation. However, because an LED driver is essentially free of output transients (except catastrophic
failures open or short), the dominant pole approach, even with reduced bandwidth, is usually the best approach.
The dominant compensation pole (ωP2) is determined by CCMP and the output resistance (RO) of the error
amplifier (typically 5 MΩ):
(17)
It may also be necessary to add one final pole at least one decade above the crossover frequency to attenuate
switching noise and, in some cases, provide better gain margin. This pole can be placed across RSNS to filter the
ESL of the sense resistor at the same time. Figure 19 shows how the compensation is physically implemented in
the system.
The high frequency pole (ωP3) can be calculated:
(18)
The total system transfer function becomes:
(19)
The resulting compensated loop gain frequency response shown in Figure 20 indicates that the system has
adequate phase margin (above 45°) if the dominant compensation pole is placed low enough, ensuring stability:
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LM3429 LM3429-Q1
1.24V
20 PA
LM3429
ROV2
ROV1
VO
OVLO
OVP
PHASE (°)
FREQUENCY (Hz)
GAIN (dB)
80
60
40
20
0
-20
-40
-60
-80
90
45
0
-45
-90
-135
-180
-225
-270
1e-1 1e1 1e3 1e5 1e7
GAIN
60° Phase Margin
PHASE
öP2
öP3
öP1
öZ1
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Feature Description (continued)
Figure 20. Compensated Loop Gain Frequency Response
7.3.8 Output Overvoltage Lockout (OVLO)
The LM3429 can be configured to detect an output (or input) overvoltage condition through the OVP pin. The pin
features a precision 1.24-V threshold with 20 µA (typical) of hysteresis current as shown in Figure 21. When the
OVLO threshold is exceeded, the GATE pin is immediately pulled low and a 20 µA current source provides
hysteresis to the lower threshold of the OVLO hysteretic band.
Figure 21. Overvoltage Protection Circuitry
If the LEDs are referenced to a potential other than ground (floating), as in the buck-boost and buck
configuration, the output voltage (VO) should be sensed and translated to ground by using a single PNP as
shown in Figure 22.
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Product Folder Links: LM3429 LM3429-Q1
1.24V
20 PA
LM3429
RUV2
RUV1
VIN
UVLO
nDIM
RUVH
(optional)
2OVHYSO RA20V xP=
¸
¸
¹
·
¨
¨
©
§
x
=
-OFFTURN 24V.1V x+
OV1 OV2
R5.0 R
1OV
R
1¸
¸
¹
·
¨
¨
©
§
x
=
-OFFTURN 24V.1V 1OV
R+2OVOV RR
LM3429
OVP
ROV2
ROV1
LED+
LED-
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Feature Description (continued)
Figure 22. Floating Output OVP Circuitry
The overvoltage turnoff threshold (VTURN-OFF) is defined as follows:
Ground Referenced
(20)
Floating
(21)
In the ground referenced configuration, the voltage across ROV2 is VO- 1.24 V whereas in the floating
configuration it is VO- 620 mV where 620 mV approximates the VBE of the PNP transistor.
The overvoltage hysteresis (VHYSO) is defined as follows:
(22)
7.3.9 Input Undervoltage Lockout (UVLO)
The nDIM pin is a dual-function input that features an accurate 1.24 V threshold with programmable hysteresis
as shown in Figure 23. This pin functions as both the PWM dimming input for the LEDs and as a VIN UVLO.
When the pin voltage rises and exceeds the 1.24 V threshold, 20 µA (typical) of current is driven out of the nDIM
pin into the resistor divider providing programmable hysteresis.
Figure 23. UVLO Circuit
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LM3429 LM3429-Q1
LM3429
RUV2
RUV1
VIN
nDIM
RUVH
QDIM
DDIM
Inverted
PWM
Standard
PWM
x
PA20
=
HYS
V¨
¨
©
§+
2UV
R¸
¸
¹
·
1UV
R
(+
x1UV
R)
2UV
R
UVH
R
2UV
RA20 x
P
HYS
V=
1¸
¸
¹
·
¨
¨
©
§
x
=
-ONTURN 24V.1V 1UV
R+2UVUV RR
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Feature Description (continued)
When using the nDIM pin for UVLO and PWM dimming concurrently, the UVLO circuit can have an extra series
resistor to set the hysteresis. This allows the standard resistor divider to have smaller resistor values minimizing
PWM delays due to a pulldown MosFET at the nDIM pin (see PWM Dimming section). In general, at least 3V of
hysteresis is necessary when PWM dimming if operating near the UVLO threshold.
The turn-on threshold (VTURN-ON) is defined as follows:
(23)
The hysteresis (VHYS) is defined as follows:
UVLO Only
(24)
PWM Dimming and UVLO
(25)
7.3.10 PWM Dimming
The active low nDIM pin can be driven with a PWM signal which controls the main NFET (Q1). The brightness of
the LEDs can be varied by modulating the duty cycle of this signal. LED brightness is approximately proportional
to the PWM signal duty cycle, so 30% duty cycle equals approximately 30% LED brightness. This function can
be ignored if PWM dimming is not required by using nDIM solely as a VIN UVLO input as described in the Input
Undervoltage Lockout (UVLO) section or by tying it directly to VCC or VIN (if less than 76VDC).
Figure 24. PWM Dimming Circuit
Figure 24 shows two ways the PWM signal can be applied to the nDIM pin:
1. Connect the dimming MosFET (QDIM) with the drain to the nDIM pin and the source to GND. Apply an
external logic-level PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn
off QDIM if no signal is present.
2. Connect the anode of a Schottky diode (DDIM) to the nDIM pin. Apply an external inverted logic-level PWM
signal to the cathode of the same diode.
A minimum on-time must be maintained in order for PWM dimming to operate in the linear region of its transfer
function. Because the controller is disabled during dimming, the PWM pulse must be long enough such that the
energy intercepted from the input is greater than or equal to the energy being put into the LEDs. For boost and
buck-boost regulators, the following condition must be maintained:
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Product Folder Links: LM3429 LM3429-Q1
tPULSE = 2 x ILED x VO X L1
VIN2
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Feature Description (continued)
(26)
In the previous equation, tPULSE is the length of the PWM pulse in seconds.
7.3.11 Startup Regulator (VCC LDO)
The LM3429 includes a high voltage, low dropout (LDO) bias regulator. When power is applied, the regulator is
enabled and sources current into an external capacitor connected to the VCC pin. The VCC output voltage is 6.9V
nominally and the supply is internally current limited to 20 mA (minimum). The recommended bypass
capacitance range for the VCC regulator is 2.2 µF to 3.3 µF. The output of the VCC regulator is monitored by an
internal UVLO circuit that protects the device during startup, normal operation, and shutdown from attempting to
operate with insufficient supply voltage.
7.3.12 Thermal Shutdown
The LM3429 includes thermal shutdown. If the die temperature reaches approximately 165°C the device will shut
down (GATE pin low), until it reaches approximately 140°C where it turns on again.
7.4 Device Functional Modes
This device has no functional modes.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 21
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,
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Inductor
The inductor (L1) is the main energy storage device in a switching regulator. Depending on the topology, energy
is stored in the inductor and transfered to the load in different ways (as an example, buck-boost operation is
detailed in the Current Regulators section). The size of the inductor, the voltage across it, and the length of the
switching subinterval (tON or tOFF) determines the inductor current ripple (ΔiL-PP). In the design process, L1 is
chosen to provide a desired ΔiL-PP. For a buck regulator the inductor has a direct connection to the load, which is
good for a current regulator. This requires little to no output capacitance therefore ΔiL-PP is basically equal to the
LED ripple current ΔiLED-PP. However, for boost and buck-boost regulators, there is always an output capacitor
which reduces ΔiLED-PP, therefore the inductor ripple can be larger than in the buck regulator case where output
capacitance is minimal or completely absent.
In general, ΔiLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED).
Therefore, for the buck regulator with no output capacitance, ΔiL-PP should also be less than 40% of ILED. For the
boost and buck-boost topologies, ΔiL-PP can be much higher depending on the output capacitance value.
However, ΔiL-PP is suggested to be less than 100% of the average inductor current (IL) to limit the RMS inductor
current.
L1 is also suggested to have an RMS current rating at least 25% higher than the calculated minimum allowable
RMS inductor current (IL-RMS).
8.1.2 LED Dynamic Resistance (rD)
When the load is a string of LEDs, the output load resistance is the LED string dynamic resistance plus RSNS.
LEDs are PN junction diodes, and their dynamic resistance shifts as their forward current changes. Dividing the
forward voltage of a single LED (VLED) by the forward current (ILED) leads to an incorrect calculation of the
dynamic resistance of a single LED (rLED). The result can be 5 to 10 times higher than the true rLED value.
Figure 25. Dynamic Resistance
Obtaining rLED is accomplished by referring to the manufacturer's LED I-V characteristic. It can be calculated as
the slope at the nominal operating point as shown in Figure 25. For any application with more than 2 series
LEDs, RSNS can be neglected allowing rDto be approximated as the number of LEDs multiplied by rLED.
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,
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Application Information (continued)
8.1.3 Output Capacitor
For boost and buck-boost regulators, the output capacitor (CO) provides energy to the load when the recirculating
diode (D1) is reverse biased during the first switching subinterval. An output capacitor in a buck topology will
simply reduce the LED current ripple (ΔiLED-PP) below the inductor current ripple (ΔiL-PP). In all cases, COis sized
to provide a desired ΔiLED-PP. As mentioned in the Inductor section, ΔiLED-PP is recommended by manufacturers to
be less than 40% of the average LED current (ILED).
COshould be carefully chosen to account for derating due to temperature and operating voltage. It must also
have the necessary RMS current rating. Ceramic capacitors are the best choice due to their high ripple current
rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested.
8.1.4 Input Capacitors
The input capacitance (CIN) provides energy during the discontinuous portions of the switching period. For buck
and buck-boost regulators, CIN provides energy during tON and during tOFF, the input voltage source charges up
CIN with the average input current (IIN). For boost regulators, CIN only needs to provide the ripple current due to
the direct connection to the inductor. CIN is selected given the maximum input voltage ripple (ΔvIN-PP) which can
be tolerated. ΔvIN-PP is suggested to be less than 10% of the input voltage (VIN).
An input capacitance at least 100% greater than the calculated CIN value is recommended to account for derating
due to temperature and operating voltage. When PWM dimming, even more capacitance can be helpful to
minimize the large current draw from the input voltage source during the rising transition of the LED current
waveform.
The chosen input capacitors must also have the necessary RMS current rating. Ceramic capacitors are again the
best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R
dieletric rating is suggested.
For most applications, TI recommends bypassing the VIN pin with an 0.1-µF ceramic capacitor placed as close as
possible to the pin. In situations where the bulk input capacitance may be far from the LM3429 device, a 10-Ω
series resistor can be placed between the bulk input capacitance and the bypass capacitor, creating a 150 kHz
filter to eliminate undesired high frequency noise coupling.
8.1.5 N-Channel MosFET (NFET)
The LM3429 requires an external NFET (Q1) as the main power MosFET for the switching regulator. Q1 is
recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe
operation during the ringing of the switch node. In practice, all switching regulators have some ringing at the
switch node due to the diode parasitic capacitance and the lead inductance. The current rating is recommended
to be at least 10% higher than the average transistor current. The power rating is then verified by calculating the
power loss given the RMS transistor current and the NFET on-resistance (RDS-ON).
In general, the NFET should be chosen to minimize total gate charge (Qg) whenever switching frequencies are
high and minimize RDS-ON otherwise. This will minimize the dominant power losses in the system. Frequently,
higher current NFETs in larger packages are chosen for better thermal performance.
8.1.6 Re-Circulating Diode
A re-circulating diode (D1) is required to carry the inductor current during tOFF. The most efficient choice for D1 is
a Schottky diode due to low forward voltage drop and near-zero reverse recovery time. Similar to Q1, D1 is
recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe
operation during the ringing of the switch node and a current rating at least 10% higher than the average diode
current. The power rating is verified by calculating the power loss through the diode. This is accomplished by
checking the typical diode forward voltage from the I-V curve on the product data sheet and multiplying by the
average diode current. In general, higher current diodes have a lower forward voltage and come in better
performing packages minimizing both power losses and temperature rise.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LM3429 LM3429-Q1
ILED
ROV1
ROV2
CO
RSNS
D1
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
AGND
RT
Q3
RUVH
PWM
RFS
CFS
COV
VIN
VCC
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
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8.2 Typical Applications
8.2.1 Basic Topology Schematics
Figure 26. Boost Regulator (VIN < VO)
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Product Folder Links: LM3429 LM3429-Q1
D1
Q2
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
ILED
ROV2
CO
RSNS
AGND
RT
Q3
RUVH
PWM
RFS
CFS
VIN
ROV1
COV
VCC
LM3429
,
LM3429-Q1
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SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
Figure 27. Buck Regulator (VIN > VO)
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM3429 LM3429-Q1
VIN
D1
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
ILED
CO
AGND
RT
Q3
RUVH
PWM
Q2
ROV2
ROV1
COV
VIN
RSNS
RFS
CFS
VIN
VCC
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
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Typical Applications (continued)
Figure 28. Buck-Boost Regulator
8.2.1.1 Design Requirements
Number of series LEDs: N
Single LED forward voltage: VLED
Single LED dynamic resistance: rLED
Nominal input voltage: VIN
Input voltage range: VIN-MAX, VIN-MIN
Switching frequency: fSW
Current sense voltage: VSNS
Average LED current: ILED
Inductor current ripple: ΔiL-PP
LED current ripple: ΔiLED-PP
Peak current limit: ILIM
Input voltage ripple: ΔvIN-PP
Output OVLO characteristics: VTURN-OFF, VHYSO
Input UVLO characteristics: VTURN-ON, VHYS
26 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
SNS
R = SNS
V
LED
I
T25
R = TSW Cf x
( )
T25
R=2
OOIN VVV -
xx 2
INTSW VCf xx
RT = 25 x VIN - VO
fSW x CT X VIN
( )
D=INO VV +
O
V
D= INO VV -
O
V
D= O
V
IN
V
rD = N x rLED
VO = N x VLED
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Operating Point
Given the number of series LEDs (N), the forward voltage (VLED) and dynamic resistance (rLED) for a single LED,
solve for the nominal output voltage (VO) and the nominal LED string dynamic resistance (rD):
(27)
(28)
Solve for the ideal nominal duty cycle (D):
Buck
(29)
Boost
(30)
Buck-boost
(31)
Using the same equations, find the minimum duty cycle (DMIN) using maximum input voltage (VIN-MAX) and the
maximum duty cycle (DMAX) using the minimum input voltage (VIN-MIN). Also, remember that D' = 1 - D.
8.2.1.2.2 Switching Frequency
Set the switching frequency (fSW) by assuming a CTvalue of 1 nF and solving for RT:
Buck (Constant Ripple vs. VIN)
(32)
Buck (Constant Ripple vs. VO)
(33)
Boost and Buck-Boost
(34)
8.2.1.2.3 Average LED Current
For all topologies, set the average LED current (ILED) knowing the desired current sense voltage (VSNS) and
solving for RSNS:
(35)
If the calculated RSNS is too far from a desired standard value, then VSNS must be adjusted to obtain a standard
value.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 27
Product Folder Links: LM3429 LM3429-Q1
IRMSCO =
-12PP-LED
iü
LED
oD LED PP
I x D
Cr x i
'
O
C = PPL
i-
'
PPLEDDSW irf8 -
'xxx
IL-RMS = 1 + 1
12 xILED
'IL-PP x D' ¸
¹
·
¨
©
§2
D'
ILED x
IL-RMS = ILED x 1 + 1
12 xILED
'IL-PP
¸
¹
·
¨
©
§2
IN
L PP SW
V x D
L1 i x f
'
IN O
L PP SW
(V V )xD
L1 i x f
'
1.24V
RHSP =RRI SNSCSHLED xx
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
Setup the suggested signal current of 100 µA by assuming RCSH = 12.4 kΩand solving for RHSP:
(36)
If the calculated RHSP is too far from a desired standard value, then RCSH can be adjusted to obtain a standard
value.
8.2.1.2.4 Inductor Ripple Current
Set the nominal inductor ripple current (ΔiL-PP) by solving for the appropriate inductor (L1):
Buck
(37)
Boost and Buck-Boost
(38)
To set the worst case inductor ripple current, use VIN-MAX and DMIN when solving for L1.
The minimum allowable inductor RMS current rating (IL-RMS) can be calculated as:
Buck
(39)
Boost and Buck-Boost
(40)
8.2.1.2.5 LED Ripple Current
Set the nominal LED ripple current (ΔiLED-PP), by solving for the output capacitance (CO):
Buck
(41)
Boost and Buck-Boost
(42)
To set the worst case LED ripple current, use DMAX when solving for CO.
The minimum allowable RMS output capacitor current rating (ICO-RMS) can be approximated:
Buck
(43)
28 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
=Dr 2
Dc
x
1Z
ZL1Dx
=Dr 2
Dc
x
1Z
ZL1
1P =Z1+D
OD Cr x
3
1P =Z2
OD Cr x
3
1P =Z1
OD Cr x
3
x
=
¨
¨
©
§+s
1Z1P ¸
¸
¹
·
0U
T
U
T¨
¨
©
§-s
1Z1Z ¸
¸
¹
·
x
=
¨
¨
©
§+s
1Z1P ¸
¸
¹
·
0U
T
U
T1
LIM
R=LIM
I
245 mV
1-DMAX
DMAX
ICO-RMS = ILED x
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
Boost and Buck-boost
(44)
8.2.1.2.6 Peak Current Limit
Set the peak current limit (ILIM) by solving for the transistor path sense resistor (RLIM):
(45)
8.2.1.2.7 Loop Compensation
Using a simple first order peak current mode control model, neglecting any output capacitor ESR dynamics, the
necessary loop compensation can be determined.
First, the uncompensated loop gain (TU) of the regulator can be approximated:
Buck
(46)
Boost and Buck-Boost
(47)
Where the pole (ωP1) is approximated:
Buck
(48)
Boost
(49)
Buck-Boost
(50)
And the RHP zero (ωZ1) is approximated:
Boost
(51)
Buck-Boost
(52)
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 29
Product Folder Links: LM3429 LM3429-Q1
x= 0U
TT -1 ¸
¸
¹
·
¨
¨
©
§s
Z1
Z
xx +1 ¸
¸
¹
·
¨
¨
©
§s
Z3P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z2P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z1P
x= 0U
TT 1
xx +1 ¸
¸
¹
·
¨
¨
©
§s
Z3P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z2P
+1 ¸
¸
¹
·
¨
¨
©
§s
Z1P
1
=
CFS 3P
10xZ
max
3P =
Z( ) 10, 1Z1P x
ZZ
1
CMP
C=6
2P
5×10
×
Ѡ
1Z1P )
,min( Z
Z
5x0U
T
2P =
Z
=
0U
T=SNSCSH RR500VD xxx
c620VD x
c
( ) LIM
LED RID1 xx+
( ) LIMHSP RRD1 xx+
=
0U
T=SNSCSH RR500VD xxx
c310VD x
c
LIMLED RI x
LIMHSP RR2 xx
SNS 620V
RR500V =
xx CSH
LIMLED RI x
0U
T=LIMHSP RR x
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
And the uncompensated DC loop gain (TU0) is approximated:
Buck
(53)
Boost
(54)
Buck-Boost
(55)
For all topologies, the primary method of compensation is to place a low-frequency dominant pole (ωP2) which
will ensure that there is ample phase margin at the crossover frequency. This is accomplished by placing a
capacitor (CCMP) from the COMP pin to GND, which is calculated according to the lower value of the pole and the
RHP zero of the system (shown as a minimizing function):
(56)
(57)
If analog dimming is used, CCMP should be approximately 4x larger to maintain stability as the LEDs are dimmed
to zero.
A high frequency compensation pole (ωP3) can be used to attenuate switching noise and provide better gain
margin. Assuming RFS = 10 Ω, CFS is calculated according to the higher value of the pole and the RHP zero of
the system (shown as a maximizing function):
(58)
(59)
The total system loop gain (T) can then be written as:
Buck
(60)
Boost and Buck-boost
(61)
8.2.1.2.8 Input Capacitance
Set the nominal input voltage ripple (ΔvIN-PP) by solving for the required capacitance (CIN):
Buck
30 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
IT-MAX = x ILED
1 - DMAX
DMAX
IT-MAX = DMAX x ILED
OMAXINMAXT VVV +
=--
O
V
=
MAXT
V-
MAXINMAXT VV -- =
1-DMAX
DMAX
ICIN-RMS = ILED x
12
ICIN-RMS = 'iL-PP
(1-DMID)DII LEDRMSCIN xx=
- MID
CIN = 'VIN-PP x fSW
ILED x D
CIN = 8 x 'VIN-PP x fSW
'iL-PP
CIN = ILED x (1 - D) x D
'VIN-PP x fSW
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,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
(62)
Boost
(63)
Buck-Boost
(64)
Use DMAX to set the worst case input voltage ripple, when solving for CIN in a buck-boost regulator and DMID = 0.5
when solving for CIN in a buck regulator.
The minimum allowable RMS input current rating (ICIN-RMS) can be approximated:
Buck
(65)
Boost
(66)
Buck-Boost
(67)
8.2.1.2.9 NFET
The NFET voltage rating should be at least 15% higher than the maximum NFET drain-to-source voltage (VT-
MAX):
Buck
(68)
Boost
(69)
Buck-Boost
(70)
The current rating should be at least 10% higher than the maximum average NFET current (IT-MAX):
Buck
(71)
Boost and Buck-Boost
(72)
Approximate the nominal RMS transistor current (IT-RMS) :
Buck
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 31
Product Folder Links: LM3429 LM3429-Q1
ROV1=R1.24V OV2
xVm620V OFFTURN -
-
ROV1 =1.24VV OFFTURN -
-
R1.24V OV2
x
ROV2 =VHYSO
A20P
FDDD VIP x=
ID-MAX = ILED
ID-MAX = (1 - DMIN) x ILED
VRD-MAX = VIN-MAX + VO
VRD-MAX = VO
VRD-MAX = VIN-MAX
DSON
2
RMSTT RIP x= -
IRMST =
-D
x
ILED
Dc
DIT- ILEDRMS x=
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
(73)
Boost and Buck-Boost
(74)
Given an NFET with on-resistance (RDS-ON), solve for the nominal power dissipation (PT):
(75)
8.2.1.2.10 Diode
The Schottky diode voltage rating should be at least 15% higher than the maximum blocking voltage (VRD-MAX):
Buck
(76)
Boost
(77)
Buck-Boost
(78)
The current rating should be at least 10% higher than the maximum average diode current (ID-MAX):
Buck
(79)
Boost and Buck-Boost
(80)
Replace DMAX with D in the ID-MAX equation to solve for the average diode current (ID). Given a diode with forward
voltage (VFD), solve for the nominal power dissipation (PD):
(81)
8.2.1.2.11 Output OVLO
For boost and buck-boost regulators, output OVLO is programmed with the turn-off threshold voltage (VTURN-OFF)
and the desired hysteresis (VHYSO). To set VHYSO, solve for ROV2:
(82)
To set VTURN-OFF, solve for ROV1:
Boost
(83)
Buck-Boost
(84)
32 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
( )
RR +x 2UV1UV
A20P
( )
xHYS A20V xP- R 2UV
R1UV
UVH
R =
RUV1 =1.24VV ONTURN -
-
R1.24V UV2
x
RUV2 =A20P
VHYS
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
A small filter capacitor (COVP = 47 pF) should be added from the OVP pin to ground to reduce coupled switching
noise.
8.2.1.2.12 Input UVLO
For all topologies, input UVLO is programmed with the turn-on threshold voltage (VTURN-ON) and the desired
hysteresis (VHYS).
Method #1: If no PWM dimming is required, a two resistor network can be used. To set VHYS, solve for RUV2:
(85)
To set VTURN-ON, solve for RUV1:
(86)
Method #2: If PWM dimming is required, a three resistor network is suggested. To set VTURN-ON, assume RUV2 =
10 kΩand solve for RUV1 as in Method #1. To set VHYS, solve for RUVH:
(87)
8.2.1.2.13 PWM Dimming Method
PWM dimming can be performed several ways:
Method #1: Connect the dimming MosFET (Q3) with the drain to the nDIM pin and the source to GND. Apply an
external PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off Q3.
Method #2: Connect the anode of a Schottky diode to the nDIM pin. Apply an external inverted PWM signal to
the cathode of the same diode.
8.2.1.2.14 Analog Dimming Method
Analog dimming can be performed several ways:
Method #1: Place a potentiometer in series with the RCSH resistor to dim the LED current from the nominal ILED
to near zero.
Method #2: Connect a controlled current source as detailed in the Analog Dimming section to the CSH pin.
Increasing the current sourced into the CSH node will decrease the LEDs from the nominal ILED to zero current.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 33
Product Folder Links: LM3429 LM3429-Q1
D1
OVP
LM3429
nDIM
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
1A
CO
RSNS
AGND
RT
RFS
CFS
ILED
Q2
ROV2
ROV1
COV
VIN
VIN
VIN
10V ± 70V
VIN
VCC
RLIM
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
8.2.2 Buck-Boost Application - 6 LEDs at 1 A
Figure 29. Buck-Boost Application - 6 LEDs at 1 A Schematic
8.2.2.1 Design Requirements
N=6
VLED = 3.5 V
rLED = 325 mΩ
VIN = 24 V
VIN-MIN = 10 V
VIN-MAX = 70 V
fSW = 700 kHz
VSNS = 100 mV
ILED = 1A
ΔiL-PP = 500 mA
ΔiLED-PP = 50 mA
ΔvIN-PP = 100 mV
ILIM = 6A
VTURN-ON = 10 V
VHYS =3V
34 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
ILED = = k0.11.24V :xA0.1=
k4.121.0 :x:
RR CSHSNS xR1.24V HSP
x
=1.24V1.24V
=
RHSP :
=k0.1
:
x
:
x0.1k12.4A1
xx RRI SNSCSHLED
:
=== 1.0
1A
RSNS ILED
mV100
VSNS
CT = 1 nF
RT = 35.7 k:
2525 =35.7 k: x 1 nF
fSW = RT x CT= 700 kHz
2525 =700 kHz x 1 nF = 35.7 k:
RT =fSW x CT
DMAX == 677.0
=
V21 V10V21 +
VO
VV IN-MINO +
21V
=21V + 70V = 0.231
DMIN = VO + VIN-MAX
VO
533.0467.01D1'D =
-
=
-
=
D== 467.0
=
V21 V24V21 +
VOVV INO +
:
=
:x
=
x
=95.1m3256rNr LEDD
V21V5.36VNV LEDO =
x
=
x
=
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,
LM3429-Q1
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SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
VTURN-OFF = 40 V
VHYSO = 10 V
8.2.2.2 Detailed Design Procedure
8.2.2.2.1 Operating Point
Solve for VOand rD:
(88)
(89)
Solve for D, D', DMAX, and DMIN:
(90)
(91)
(92)
(93)
8.2.2.2.2 Switching Frequency
Assume CT= 1 nF and solve for RT:
(94)
The closest standard resistor is actually 35.7 kΩtherefore the fSW is:
(95)
The chosen components from step 2 are:
(96)
8.2.2.2.3 Average LED Current
Solve for RSNS:
(97)
Assume RCSH = 12.4 kΩand solve for RHSP:
(98)
The closest standard resistor for RSNS is actually 0.1Ωand for RHSP is actually 1 kΩtherefore ILED is:
(99)
The chosen components from step 3 are:
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 35
Product Folder Links: LM3429 LM3429-Q1
F8.6COP=
x
A1
=ILED
IRMSCO- =
1- 0.677
677.0 1.45A
x1- DMAX
DMAX =
DILED x
=
'iPP-LED SW
fxrDxCO
0
= =
kHz00795.1 xx:5 mA
467.0A1 x F8.6 P
'iPP-LED
f
'i
rSWPP-LEDD xx DILED x
CO=
0
= F84.6 P=
kHz007mA595.1 xx:
467.0A1 x
CO
H331L P
=
A88.1
I
12
1
1
I
RMSL
RMSL
=
+
x
=
-
-
I
ILED
RMSL x
=
-12
1
12
x
+¸
¸
¹
·
¨
¨
©
§Di PPL c
x
'-
ILED
Dc
533.0mA485 2
x
x¸
¸
¹
·
¨
¨
©
§A1
533.0 A1
PP- ==
LDVIN xf1L SW
xkHz007H33 xP
467.0V42 x mA485
=
'i
== DVIN xfSW
x467.0V42 x PH
32
=
1L PP-
'iLkHz007500 mA x
:
k4.21RCSH =
:
k1RR HSN ==
HSP
0.1R NSS =:
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
(100)
8.2.2.2.4 Inductor Ripple Current
Solve for L1:
(101)
The closest standard inductor is 33 µH therefore the actual ΔiL-PP is:
(102)
Determine minimum allowable RMS current rating:
(103)
The chosen component from step 4 is:
(104)
8.2.2.2.5 Output Capacitance
Solve for CO:
(105)
The closest standard capacitor is 6.8 µF therefore the actual ΔiLED-PP is:
(106)
Determine minimum allowable RMS current rating:
(107)
The chosen components from step 5 are:
(108)
36 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
F1.0CFS P=
F0.22CCOMP P=
:10RFS =
1
= F091.0
1P==CFS 10:xsec
rad
M1.1
10:3P
Zx
1010max 1P xZ
=
x
=,1Z1P ZZ
sec
rad
k110
=sec
rad
M1.110=
x
3P
Z
3P
Z
F17.0
1
1
CCMP µ===
173.1 sec
rad 6
×Ω
5 10× 6
2P ×ѠΩ5 10×
= = sec
rad
173.1=
sec
rad
k37
56305x56305x 1Z
Z
2P =Z),min( 1Z1P ZZ
T5 0U
x
=T 0U = 5630=
04.0A1467.1 :
xx V620533.0 x
( )
D1+RI LIMLED xx V620D x
c
sec
rad
k37=== 533.095.1 2
x:H33467.0 Px
Dr 2
Dc
xL1Dx
1Z
Z
=sec
rad
= 110k
1 + D
ZP1 = 1.467
1.95: x 6.8 PF
rD x CO
0.04:RLIM =
:04.0 === 6.13A
mV245mV245
ILIM RLIM
:=== 041.0
6A
RLIM mV245mV245
ILIM
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,
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SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
8.2.2.2.6 Peak Current Limit
Solve for RLIM:
(109)
The closest standard resistor is 0.04 Ωtherefore ILIM is:
(110)
The chosen component from step 6 is:
(111)
8.2.2.2.7 Loop Compensation
ωP1 is approximated:
(112)
ωZ1 is approximated:
(113)
TU0 is approximated:
(114)
To ensure stability, calculate ωP2:
(115)
Solve for CCMP:
(116)
To attenuate switching noise, calculate ωP3:
(117)
Assume RFS = 10 Ωand solve for CFS:
(118)
The chosen components from step 7 are:
(119)
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 37
Product Folder Links: LM3429 LM3429-Q1
D1 o 12A, 100V, DPAK
mW600mV600A1VIP FDDD =
x
=
x
=
A1II LEDMAXD ==
-
V91V21V70VVV OMAXINMAXRD =
+
=
+
=--
Q1 o 32A, 100V, DPAK
mW82m50A28.1RIP 2
DSON
2
RMSTT =
:x
=
x
=-
x
IRMST =
-ILED
Dc=xA28.1
=
0.467
A1
533.0
D
=A2.1A1 =
x
677.01- 677.0
IMAXT-
V91V21V70VVV OMAXINMAXT =+=+= --
CIN = 3 x 4.7 PF
x
A1
=ILED
IRMSIN- =
1- 0.677
677.0 1.45A
x1- DMAX
DMAX =
CIN == kHz700mV100 x 467.0A1 x F66.6 P=
f
'vSWPPIN- x
DILED x
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
8.2.2.2.8 Input Capacitance
Solve for the minimum CIN:
(120)
To minimize power supply interaction a 200% larger capacitance of approximately 14 µF is used, therefore the
actual ΔvIN-PP is much lower. Because high voltage ceramic capacitor selection is limited, three 4.7 µF X7R
capacitors are chosen.
Determine minimum allowable RMS current rating:
(121)
The chosen components from step 8 are:
(122)
8.2.2.2.9 NFET
Determine minimum Q1 voltage rating and current rating:
(123)
(124)
A 100-V NFET is chosen with a current rating of 32A due to the low RDS-ON = 50 mΩ. Determine IT-RMS and PT:
(125)
(126)
The chosen component from step 9 is:
(127)
8.2.2.2.10 Diode
Determine minimum D1 voltage rating and current rating:
(128)
(129)
A 100-V diode is chosen with a current rating of 12 A and VDF = 600 mV. Determine PD:
(130)
The chosen component from step 10 is:
(131)
8.2.2.2.11 Input UVLO
Solve for RUV2:
38 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
ROV1 = 15.8 k:
ROV2 = 499 k:
= 39.8V
1.24V x (0.5 x 15.8 k: + 499 k:)
VTURN-OFF = 15.8 k:
1.24V x (0.5 x ROV1 + ROV2)
VTURN-OFF = ROV1
:== k15.7
-
-0.62VV OFFTURN
xR1.24V OV2
=ROV1
:x k4991.24V- 0.62V40V
29.98VA20k499A20RV OVHYSO =
x:
=
x
=P
P
=== A
20
10V
ROV2 P:k500
VHYSO
A20P
k:501R 2UV =k:12R 1UV =
RUV1
()
RR1.24V UV2UV1+x
VONTURN =
-
= V10.1=
( )
k150k211.24V :+:xk21 :
VONTURN-
:== k2.21
-
-1.24VV ONTURN
xR1.24V UV2
=RUV1
:x k1501.24V-1.24V10V
3VA20k150A20RV 2UVHYS =
x:
=
x
=PP
=== A
20
3V
RUV2 P:k150
VHYS
A20P
LM3429
,
LM3429-Q1
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SNVS616H –APRIL 2009–REVISED JULY 2015
Typical Applications (continued)
(132)
The closest standard resistor is 150 kΩtherefore VHYS is:
(133)
Solve for RUV1:
(134)
The closest standard resistor is 21 kΩmaking VTURN-ON:
(135)
The chosen components from step 11 are:
(136)
8.2.2.2.12 Output OVLO
Solve for ROV2:
(137)
The closest standard resistor is 499 kΩtherefore VHYSO is:
(138)
Solve for ROV1:
(139)
The closest standard resistor is 15.8 kΩmaking VTURN-OFF:
(140)
The chosen components from step 12 are:
(141)
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 39
Product Folder Links: LM3429 LM3429-Q1
VIN (V)
EFFICIENCY (%)
100
95
90
85
80
75
70 0 16 32 48 64 80
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Typical Applications (continued)
Table 1. Design 1 Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 0.22 µF X7R 10% 25 V MURATA GRM21BR71E224KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
3 CIN 4.7 µF X7R 10% 100 V TDK C5750X7R2A475K
1 CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF
1 L1 33 µH 20% 6.3 A COILCRAFT MSS1278-333MLB
1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682
1 Q2 PNP 150 V 600 m A FAIRCHILD MMBT5401
1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
1 RLIM 0.04 Ω1% 1W VISHAY WSL2512R0400FEA
1 ROV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.1 Ω1% 1W VISHAY WSL2512R1000FEA
1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA
1 RUV1 21 kΩ1% VISHAY CRCW080521K0FKEA
1 RUV2 150 kΩ1% VISHAY CRCW0805150KFKEA
8.2.2.3 Application Curve
Figure 30. Buck-Boost Efficiency vs Input Voltage, VO= 6 LEDs
40 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
CO
D1
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
AGND
RT
Q2
RUVH
PWM
RSNS
RFS
CFS
ILED
ROV1
ROV2
COV
VIN
8V - 28V
1A
VCC
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
8.2.3 Boost PWM Dimming Application - 9 LEDs at 1 A
Figure 31. Boost PWM Dimming Application - 9 LEDs at 1 A Schematic
8.2.3.1 Detailed Design Procedure
Table 2. Design 2 Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
2 CCMP, CFS 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
2, 1 CIN, CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 60 V 5 A COMCHIP CDBC560-G
1 L1 33 µH 20% 6.3 A COILCRAFT MSS1278-333MLB
1 Q1 NMOS 60 V 8 A VISHAY SI4436DY
1 Q2 NMOS 60 V 115 mA ON SEMI 2N7002ET1G
2 RCSH, ROV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
1 RLIM 0.06 Ω1% 1 W VISHAY WSL2512R0600FEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.1 Ω1% 1 W VISHAY WSL2512R1000FEA
1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA
1 RUV1 1.82 kΩ1% VISHAY CRCW08051K82FKEA
1 RUV2 10 kΩ1% VISHAY CRCW080510KFKEA
1 RUVH 17.8 kΩ1% VISHAY CRCW080517K8FKEA
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 41
Product Folder Links: LM3429 LM3429-Q1
D1
OVP
LM3429
nDIM
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
2A
CO
RSNS
AGND
RT
RFS
CFS
RADJ
ILED
Q2
ROV2
ROV1
COV
VIN
VIN
VIN
10V ± 30V
VIN
VCC
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
8.2.4 Buck-Boost Analog Dimming Application - 4 LEDs at 2A
Figure 32. Buck-Boost Analog Dimming Application - 4 LEDs at 2 A Schematic
8.2.4.1 Detailed Design Procedure
Table 3. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 1 µF X7R 10% 10 V MURATA GRM21BR71A105KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 50 V MURATA GRM21BR71E104KA01L
2, 1 CIN, CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 60 V 5 A VISHAY CDBC560-G
1 L1 22 µH 20% 7.2 A COILCRAFT MSS1278-223MLB
1 Q1 NMOS 60 V 8 A VISHAY SI4436DY
1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401
1 RADJ 1-MΩpotentiometer BOURNS 3352P-1-105
1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA
1 ROV1 18.2 kΩ1% VISHAY CRCW080518K2FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.05 Ω1% 1 W VISHAY WSL2512R0500FEA
1 RT41.2 kΩ1% VISHAY CRCW080541K2FKEA
42 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Table 3. Bill of Materials (continued)
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 RUV1 21 kΩ1% VISHAY CRCW080521K0FKEA
1 RUV2 150 kΩ1% VISHAY CRCW0805150KFKEA
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 43
Product Folder Links: LM3429 LM3429-Q1
700 mA
CO
RSNS
D1
OVP
LM3429
nDIM
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
AGND
RT
RFS
CFS
RBIAS
RMAX
Q2
Q3
VREF
RADJ
Q4
RCSH ILED
ROV1
ROV2
COV
VIN
18V - 38V
VIN
VCC
VIN
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
8.2.5 Boost Analog Dimming Application - 12 LEDs at 700 mA
Figure 33. Boost Analog Dimming Application - 12 LEDs at 700 mA Schematic
8.2.5.1 Detailed Design Procedure
Table 4. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 1 µF X7R 10% 10 V MURATA GRM21BR71A105KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 50 V MURATA GRM21BR71E104KA01L
2, 1 CIN, CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF
1 L1 47 µH 20% 5.3 A COILCRAFT MSS1278-473MLB
1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682
1 Q2 NPN 40 V 200 mA FAIRCHILD MMBT3904
1 Q3, Q4 (dual pack) Dual PNP 40 V 200 mA FAIRCHILD FFB3906
1 RADJ 100 kΩpotentiometer BOURNS 3352P-1-104
1 RBIAS 40.2 kΩ1% VISHAY CRCW080540K2FKEA
1 RCSH, ROV1, RUV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1.05 kΩ1% VISHAY CRCW08051K05FKEA
1 RLIM 0.06 Ω1% 1 W VISHAY WSL2512R0600FEA
1 RMAX 4.99 kΩ1% VISHAY CRCW08054K99FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.15 Ω1% 1 W VISHAY WSL2512R1500FEA
44 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Table 4. Bill of Materials (continued)
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA
1 RUV2 100 kΩ1% VISHAY CRCW0805100KFKEA
1 VREF 5 V precision reference TI LM4040
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 45
Product Folder Links: LM3429 LM3429-Q1
D1
OVP
LM3429
nDIM
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
500 mA
CO
RSNS
AGND
RT
RFS
CFS
ILED
Q2
ROV2
ROV1
COV
VIN
VIN
VIN
10V ± 70V
VIN
VCC
RUVH
PWM
D2
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
8.2.6 Buck-Boost PWM Dimming Application - 6 LEDs at 500 mA
Figure 34. Buck-Boost PWM Dimming Application - 6 LEDs at 500 mA
8.2.6.1 Detailed Design Procedure
Table 5. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 0.68 µF X7R 10% 25 V MURATA GRM21BR71E684KA88L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
3 CIN 4.7 µF X7R 10% 100 V TDK C5750X7R2A475K
1 CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF
1 D2 Schottky 30 V 500 mA ON SEMI BAT54T1G
1 L1 68 µH 20% 4.3 A COILCRAFT MSS1278-683MLB
1 Q1 NMOS 100 V 32 A VISHAY FDD3682
1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401
1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
1 ROV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.2 Ω1% 1 W VISHAY WSL2512R2000FEA
1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA
1 RUV1 1.43 kΩ1% VISHAY CRCW08051K43FKEA
1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA
46 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Table 5. Bill of Materials (continued)
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 RUVH 17.4 kΩ1% VISHAY CRCW080517K4FKEA
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 47
Product Folder Links: LM3429 LM3429-Q1
D1
Q2
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
ILED
ROV2
CO
RSNS
AGND
RT
RFS
CFS
VIN
ROV1
COV
VCC
1.25A
15V ± 50V
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
8.2.7 Buck Application - 3 LEDS at 1.25 A
Figure 35. Buck Application - 3 LEDS at 1.25 A Schematic
8.2.7.1 Detailed Design Procedure
Table 6. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 0.015 µF X7R 10% 50 V MURATA GRM21BR71H153KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.01 µF X7R 10% 50 V MURATA GRM21BR71H103KA01L
2 CIN 6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 CO1 µF X7R 10% 50 V TDK C4532X7R1H105K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 60V 5 A COMCHIP CDBC560-G
1 L1 22 µH 20% 7.3 A COILCRAFT MSS1278-223MLB
1 Q1 NMOS 60 V 8 A VISHAY SI4436DY
1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401
1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA
1 ROV1 21.5 kΩ1% VISHAY CRCW080521K5FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.08 Ω1% 1 W VISHAY WSL2512R0800FEA
48 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
Table 6. Bill of Materials (continued)
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 RUV1 11.5 kΩ1% VISHAY CRCW080511K5FKEA
1 RUV2 100 kΩ1% VISHAY CRCW0805100KFKEA
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 49
Product Folder Links: LM3429 LM3429-Q1
D1
OVP
LM3429
nDIM
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
Q1
CCMP
RCSH CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP 2.5A
CO
RSNS
AGND
RT
RFS
CFS
ILED
Q2
ROV2
ROV1
COV
VIN
VIN
VIN
15V ± 60V
VIN
VCC
VREF
RBIAS D2
RNTC
RGAIN
VIN
RLIM
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
8.2.8 Buck-Boost Thermal Foldback Application - 8 LEDs at 2.5 A
Figure 36. Buck-Boost Thermal Foldback Application - 8 LEDs at 2.5 A Schematic
8.2.8.1 Detailed Design Procedure
Table 7. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
3 CIN 4.7 µF X7R 10% 100 V TDK C5750X7R2A475K
1 CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF
1 L1 22 µH 20% 7.2 A COILCRAFT MSS1278-223MLB
1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682
1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401
2 RCSH, ROV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA
2 RLIM, RSNS 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA
1 RUV1 13.7 kΩ1% VISHAY CRCW080513K7FKEA
1 RUV2 150 kΩ1% VISHAY CRCW0805150KFKEA
50 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
CO
D1
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
AGND
RT
RSNS
RFS
CFS
ILED
ROV1
ROV2
COV
VIN
9V - 36V
750 mA
VCC
L2
CSEP
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
8.2.9 SEPIC Application - 5 LEDs at 750 mA
Figure 37. 5 LEDs at 750 mA
8.2.9.1 Detailed Design Procedure
Table 8. Bill of Materials
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 LM3429 Boost controller TI LM3429MH
1 CCMP 0.47 µF X7R 10% 25 V MURATA GRM21BR71E474KA01L
1 CF2.2 µF X7R 10% 16 V MURATA GRM21BR71C225KA12L
1 CFS 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L
2, 1 CIN, CO6.8 µF X7R 10% 50 V TDK C4532X7R1H685K
1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A
1 CSEP 1 µF X7R 10% 100 V TDK C4532X7R2A105K
1 CT1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D
1 D1 Schottky 60 V 5 A COMCHIP CDBC560-G
1 L1, L2 68 µH 20% 4.3 A COILCRAFT DO3340P-683
1 Q1 NMOS 60 V 8 A VISHAY SI4436DY
1 Q2 NMOS 60 V 115 mA ON SEMI 2N7002ET1G
1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA
1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA
2 RHSP, RHSN 750 Ω1% VISHAY CRCW0805750RFKEA
1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA
2 ROV1, RUV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA
1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA
1 RSNS 0.1 Ω1% 1 W VISHAY WSL2512R1000FEA
1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 51
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
Table 8. Bill of Materials (continued)
QTY PART ID PART VALUE MANUFACTURER PART NUMBER
1 RUV2 100 kΩ1% VISHAY CRCW0805100KFKEA
52 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
9 Power Supply Recommendations
The device is designed to operate from an input voltage supply range from 4.5 V to 75 V. This input supply
should be well regulated. If the input supply is located more than a few inches from the EVM or PCB, additional
bulk capacitance may be required in addition to the ceramic bypass capacitors.
9.1 Input Supply Current Limit
It is important to set the output current limit of your input supply to an appropriate value to avoid delays in your
converter analysis and optimization. If not set high enough, current limit can be tripped during start-up or when
your converter output power is increased, causing a foldback or shut-down condition. It is a common oversight
when powering up a converter for the first time.
10 Layout
10.1 Layout Guidelines
The performance of any switching regulator depends as much upon the layout of the PCB as the component
selection. Following a few simple guidelines will maximimize noise rejection and minimize the generation of EMI
within the circuit.
Discontinuous currents are the most likely to generate EMI; therefore, take care when routing these paths. The
main path for discontinuous current in the LM3429 buck regulator contains the input capacitor (CIN), the
recirculating diode (D1), the N-channel MosFET (Q1), and the switch sense resistor (RLIM). In the LM3429 boost
and buck-boost regulators, the discontinuous current flows through the output capacitor (CO), D1, Q1, and RLIM.
In either case, this loop should be kept as small as possible and the connections between all the components
should be short and thick to minimize parasitic inductance. In particular, the switch node (where L1, D1 and Q1
connect) should be just large enough to connect the components. To minimize excessive heating, large copper
pours can be placed adjacent to the short current path of the switch node.
The RCT, COMP, CSH, IS, HSP and HSN pins are all high-impedance inputs which couple external noise easily,
therefore the loops containing these nodes should be minimized whenever possible.
In some applications the LED or LED array can be far away (several inches or more) from the LM3429, or on a
separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or
separated from the rest of the regulator, the output capacitor should be placed close to the LEDs to reduce the
effects of parasitic inductance on the AC impedance of the capacitor.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 53
Product Folder Links: LM3429 LM3429-Q1
Note critical paths and component placement:
Minimize power loop containing discontinuous currents
Minimize signal current loops (components close to IC)
x Ground plane under IC for signal routing helps minimize noise coupling
ILED
ROV1
ROV2
CO
RSNS
D1
OVP
LM3429
nDIM
VIN
PGND
NC
DAP
GATE
COMP
CSH
RCT
IS
HSN
HSP
L1
CIN
CBYP
RLIM
Q1
CCMP
RCSH
CT
RUV2
RUV1
1
2
3
4
5
6
7
14
13
12
11
10
9
8
RHSN
RHSP
AGND
RT
Q3
RUVH
PWM
RFS
CFS
COV
VIN
VCC
Input
Power
GND
STAR GROUND
Power Ground
discontinuous switching
frequency currents
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
10.2 Layout Example
Figure 38. LM3429 Layout Guideline
54 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
www.ti.com
SNVS616H –APRIL 2009–REVISED JULY 2015
11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
•AN-1986 LM3429 Boost Evaluation Board,SNVA404
•AN-1985 LM3429 Buck-Boost Evaluation Board,SNVA403
11.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 9. Related Links
TECHNICAL TOOLS & SUPPORT &
PARTS PRODUCT FOLDER SAMPLE & BUY DOCUMENTS SOFTWARE COMMUNITY
LM3429 Click here Click here Click here Click here Click here
LM3429-Q1 Click here Click here Click here Click here Click here
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.5 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
Copyright © 2009–2015, Texas Instruments Incorporated Submit Documentation Feedback 55
Product Folder Links: LM3429 LM3429-Q1
LM3429
,
LM3429-Q1
SNVS616H –APRIL 2009–REVISED JULY 2015
www.ti.com
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 —TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
56 Submit Documentation Feedback Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: LM3429 LM3429-Q1
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM3429MH/NOPB ACTIVE HTSSOP PWP 14 94 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429
MH
LM3429MHX/NOPB ACTIVE HTSSOP PWP 14 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429
MH
LM3429Q1MH/NOPB ACTIVE HTSSOP PWP 14 94 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429
Q1MH
LM3429Q1MHX/NOPB ACTIVE HTSSOP PWP 14 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429
Q1MH
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 2
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM3429, LM3429-Q1 :
•Catalog: LM3429
•Automotive: LM3429-Q1
NOTE: Qualified Version Definitions:
•Catalog - TI's standard catalog product
•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM3429MHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
LM3429Q1MHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 6-Nov-2015
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM3429MHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0
LM3429Q1MHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 6-Nov-2015
Pack Materials-Page 2
MECHANICAL DATA
PWP0014A
www.ti.com
MXA14A (Rev A)
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