LT8302MPS8E#PBF - LINEAR TECHNOLOGY

LT8302

8302fa

For more information www.linear.com/LT8302

Typical applicaTion

FeaTures DescripTion

42VIN Micropower No-Opto

Isolated Flyback Converter

with 65V/3.6A Switch

The LT

8302 is a monolithic micropower isolated flyback

converter. By sampling the isolated output voltage directly

from the primary-side flyback waveform, the part requires

no third winding or opto-isolator for regulation. The output

voltage is programmed with two external resistors and a

third optional temperature compensation resistor. Bound-

ary mode operation provides a small magnetic solution with

excellent load regulation. Low ripple Burst Mode operation

maintains high efficiency at light load while minimizing the

output voltage ripple. A 3.6A, 65V DMOS power switch

is integrated along with all the high voltage circuitry and

control logic into a thermally enhanced 8-lead SO package.

The LT8302 operates from an input voltage range of 2.8V

to 42V and delivers up to 18W of isolated output power.

The high level of integration and the use of boundary

and low ripple burst modes result in a simple to use, low

component count, and high efficiency application solution

for isolated power delivery.

2.8V to 32VIN/5VOUT Isolated Flyback Converter

applicaTions

n 2.8V to 42V Input Voltage Range

n 3.6A, 65V Internal DMOS Power Switch

n Low Quiescent Current:

106µA in Sleep Mode

380µA in Active Mode

n Quasi-Resonant Boundary Mode Operation at

Heavy Load

n Low Ripple Burst Mode

Operation at Light Load

n Minimum Load < 0.5% (Typ) of Full Output

n No Transformer Third Winding or Opto-Isolator

Required for Output Voltage Regulation

n Accurate EN/UVLO Threshold and Hysteresis

n Internal Compensation and Soft-Start

n Temperature Compensation for Output Diode

n Output Short-Circuit Protection

n Thermally Enhanced 8-Lead SO Package

n Isolated Automotive, Industrial, Medical Power

Supplies

n Isolated Auxiliary/Housekeeping Power Supplies

L, LT, LTC , LT M, Linear Technology, the Linear logo and Burst Mode are registered trademarks

of Linear Technology Corporation. All other trademarks are the property of their respective

owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522.

Efficiency vs Load Current

VIN

LT8302

9µH

VIN

2.8V TO 32V 3:1

1µH

RFB

RREF

EN/UVLO

470pF

10µF

1µF

220µF

10mA TO 1.1A (VIN = 5V)

10mA TO 2.0A (VIN = 12V)

10mA TO 2.9A (VIN = 24V)

VOUT–

39Ω

154k

•

115k 10k

8302 TA01a

GND

INTVCC

VOUT+

LOAD CURRENT (A)

EFFICIENCY (%)

1.5 2.5

8302 TA01b

0.5 1.0 2.0 3.0

VIN = 5V

VIN = 12V

VIN = 24V

FRONT PAGE APPLICATION

LT8302

8302fa

For more information www.linear.com/LT8302

pin conFiguraTionabsoluTe MaxiMuM raTings

SW (Note 2) ..............................................................65V

VIN ............................................................................42V

EN/UVLO ....................................................................VIN

RFB ........................................................VIN – 0.5V to VIN

Current Into RFB ....................................................200µA

INTVCC, RREF, TC .........................................................4V

Operating Junction Temperature Range (Notes 3, 4)

LT8302E, LT8302I.............................. –40°C to 125°C

LT8302H ............................................–40°C to 150°C

LT8302MP ......................................... –55°C to 150°C

Storage Temperature Range .................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec) ...................300°C

(Note 1)

TOP VIEW

RREF

RFB

EN/UVLO

INTVCC

VIN

GND

S8E PACKAGE

8-LEAD PLASTIC SO

GND

θJA = 33°C/W

EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE

LT8302ES8E#PBF LT8302ES8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C

LT8302IS8E#PBF LT8302IS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C

LT8302HS8E#PBF LT8302HS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C

LT8302MPS8E#PBF LT8302MPS8E#TRPBF 8302 8-Lead Plastic SO –55°C to 150°C

Consult LTC Marketing for parts specified with wider operating temperature ranges.

Consult LTC Marketing for information on nonstandard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

LT8302

8302fa

For more information www.linear.com/LT8302

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT

VIN VIN Voltage Range l2.8 42 V

IQVIN Quiescent Current VEN/UVLO = 0.3V

VEN/UVLO = 1.1V

Sleep Mode (Switch Off)

Active Mode (Switch On)

0.5

106

380

2 µA

µA

EN/UVLO Shutdown Threshold For Lowest Off IQl0.3 0.75 V

EN/UVLO Enable Threshold Falling l1.178 1.214 1.250 V

EN/UVLO Enable Hysteresis 14 mV

IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V

VEN/UVLO = 1.1V

VEN/UVLO = 1.3V

–0.1

2.3

–0.1

2.5

0.1

2.7

0.1

µA

VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.85 3 3.1 V

IINTVCC INTVCC Current Limit VINTVCC = 2.8V 10 13 16 mA

INTVCC UVLO Threshold Falling 2.39 2.47 2.55 V

INTVCC UVLO Hysteresis 105 mV

(RFB – VIN) Voltage IRFB = 75µA to 125µA –50 50 mV

RREF Regulation Voltage l0.98 1.00 1.02 V

RREF Regulation Voltage Line Regulation 2.8V ≤ VIN ≤ 42V –0.01 0 0.01 %/V

VTC TC Pin Voltage 1.00 V

ITC TC Pin Current VTC = 1.2V

VTC = 0.8V 12 15

–200 18 µA

µA

fMIN Minimum Switching Frequency 11.3 12 12.7 kHz

tON(MIN) Minimum Switch-On Time 160 ns

tOFF(MAX) Maximum Switch-Off Time Backup Timer 170 µs

ISW(MAX) Maximum Switch Current Limit 3.6 4.5 5.4 A

ISW(MIN) Minimum Switch Current Limit 0.78 0.87 0.96 A

RDS(ON) Switch On-Resistance ISW = 1.5A 80 mΩ

ILKG Switch Leakage Current VSW = 65V 0.1 0.5 µA

tSS Soft-Start Timer 11 ms

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The SW pin is rated to 65V for transients. Depending on the

leakage inductance voltage spike, operating waveforms of the SW pin

should be derated to keep the flyback voltage spike below 65V as shown

in Figure 5.

Note 3: The LT8302E is guaranteed to meet performance specifications

from 0°C to 125°C junction temperature. Specifications over the –40°C

to 125°C operating junction temperature range are assured by design,

characterization and correlation with statistical process controls. The

LT8302I is guaranteed over the full –40°C to 125°C operating junction

temperature range. The LT8302H is guaranteed over the full –40°C to

150°C operating junction temperature range. The LT8302MP is guaranteed

over the full –55°C to 150°C operating junction temperature range. High

junction temperatures degrade operating lifetimes. Operating lifetime is

derated at junction temperature greater than 125°C.

Note 4: The LT8302 includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction

temperature will exceed 150°C when overtemperature protection is active.

Continuous operation above the specified maximum operating junction

temperature may impair device reliability.

LT8302

8302fa

For more information www.linear.com/LT8302

Typical perForMance characTerisTics

Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms

VIN Shutdown Current

VIN Quiescent Current,

Sleep Mode

VIN Quiescent Current,

Active Mode

Output Load and Line Regulation Output Temperature Variation

Switching Frequency

vs Load Current

TA = 25°C, unless otherwise noted.

TEMPERATURE (°C)

–50

OUTPUT VOLTAGE (V)

5.1

5.2

5.3

25 75

8302 G02

5.0

4.9

–25 0 50 100 150125

4.8

4.7

FRONT PAGE APPLICATION

VIN = 12V

IOUT = 1A

RTC = 115k

RTC = OPEN

VSW

20V/DIV

VOUT

50mV/DIV

2µs/DIV

FRONT PAGE APPLICATION

VIN = 12V

IOUT = 2A

8302 G04

VSW

20V/DIV

VOUT

50mV/DIV

2µs/DIV

FRONT PAGE APPLICATION

VIN = 12V

IOUT = 0.5A

8302 G05

VSW

20V/DIV

VOUT

50mV/DIV

20µs/DIV

FRONT PAGE APPLICATION

VIN = 12V

IOUT = 10mA

8302 G06

VIN (V)

IQ (µA)

8302 G07

010 20 30 50

TJ = 150°C

TJ = 25°C

TJ = –55°C

VIN (V)

IQ (µA)

100

110

120

130

140

10 20 30 40

8302 G08

TJ = 150°C

TJ = –55°C

TJ = 25°C

VIN (V)

IQ (µA)

380

400

420

8302 G09

360

340

320 10 20 30 50

TJ = 150°C

TJ = –55°C

TJ = 25°C

LOAD CURRENT (A)

0 0.5

FREQUENCY (kHz)

200

500

1.0 2.0 2.5

8302 G03

100

400

300

1.5 3.0

VIN = 5V

VIN = 12V

VIN = 24V

FRONT PAGE APPLICATION

LOAD CURRENT (A)

OUTPUT VOLTAGE (V)

5.15

1.5

8302 G01

5.00

4.90

0.5 1.0 2.0

4.85

4.80

5.20

5.10

5.05

4.95

2.5 3.0

VIN = 5V

VIN = 12V

VIN = 24V

LT8302

8302fa

For more information www.linear.com/LT8302

Typical perForMance characTerisTics

INTVCC Voltage vs VIN INTVCC UVLO Threshold (RFB-VIN) Voltage

RREF Regulation Voltage RREF Line Regulation TC Pin Voltage

EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature

TA = 25°C, unless otherwise noted.

TEMPERATURE (°C)

–50

VEN/UVLO (V)

1.220

1.230

150

8302 G10

1.210

1.200 050 100

–25 25 75 125

1.240

1.215

1.225

1.205

1.235 RISING

FALLING

TEMPERATURE (°C)

–50

IHYST (µA)

25 75 150

8302 G11

0–25 0 50 100 125

TEMPERATURE (°C)

–50

2.80

VINTVCC (V)

2.85

2.90

2.95

3.00

0 50 100 150

8302 G12

3.05

3.10

–25 25 75 125

IINTVCC = 0mA

IINTVCC = 10mA

VIN (V)

VINTVCC (V)

2.95

3.00

3.05

35 4020 25 30

8302 G13

2.90

2.85

10 15 45

2.80

3.10

IINTVCC = 0mA

IINTVCC = 10mA

TEMPERATURE (°C)

–50

2.2

VINTVCC (V)

2.3

2.4

2.5

2.6

0 50 100 150

8302 G14

2.7

2.8

–25 25 75 125

FALLING

RISING

TEMPERATURE (°C)

–50

VOLTAGE (mV)

150

8302 G15

–20

–40 050 100

–25 25 75 125

–10

–30

30 IRFB = 125µA

IRFB = 100µA

IRFB = 75µA

TEMPERATURE (°C)

–50

0.990

VRREF (V)

0.992

0.996

0.998

1.000

1.010

1.004

050 75

8302 G16

0.994

1.006

1.008

1.002

–25 25 100 125 150

VIN (V)

VRREF (V)

1.002

1.006

1.010

8302 G17

0.998

0.994

1.000

1.004

1.008

0.996

0.992

0.990 10 20 30 50

TEMPERATURE (°C)

–50

VTC (V)

1.1

1.3

150

8302 G18

0.9

0.7 050 100

–25 25 75 125

1.5

1.0

1.2

0.8

1.4

LT8302

8302fa

For more information www.linear.com/LT8302

Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time

RDS(ON) Switch Current Limit Maximum Switching Frequency

Typical perForMance characTerisTics

TA = 25°C, unless otherwise noted.

TEMPERATURE (°C)

–50

RESISTANCE (mΩ)

120

160

200

25 75 150

8302 G19

0–25 0 50 100 125

TEMPERATURE (°C)

–50

ISW (A)

25 75 150

8302 G20

0–25 0 50 100 125

MAXIMUM CURRENT LIMIT

MINIMUM CURRENT LIMIT

TEMPERATURE (°C)

–50

FREQUENCY (kHz)

300

400

500

25 75 150

8302 G21

200

100

0–25 0 50 100 125

TEMPERATURE (°C)

–50

FREQUENCY (kHz)

25 75 150

8302 G22

0–25 0 50 100 125

TEMPERATURE (°C)

–50

TIME (ns)

100

200

300

400

–25 0 25 50

8302 G23

75 100 125 150

TEMPERATURE (°C)

–50

TIME (ns)

100

200

300

400

–25 0 25 50

8302 G24

75 100 125 150

LT8302

8302fa

For more information www.linear.com/LT8302

pin FuncTions

EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The

EN/UVLO pin is used to enable the LT8302. Pull the pin

below 0.3V to shut down the LT8302. This pin has an ac-

curate 1.214V threshold and can be used to program a VIN

undervoltage lockout (UVLO) threshold using a resistor

divider from VIN to ground. A 2.5µA current hysteresis

allows the programming of VIN UVLO hysteresis. If neither

function is used, tie this pin directly to VIN.

INTVCC (Pin 2): Internal 3V Linear Regulator Output. The

INTVCC pin is supplied from VIN and powers the internal

control circuitry and gate driver. Do not overdrive the

INTVCC pin with any external supply, such as a third winding

supply. Locally bypass this pin to ground with a minimum

1µF ceramic capacitor.

VIN (Pin 3): Input Supply. The VIN pin supplies current to

the internal circuitry and serves as a reference voltage for

the feedback circuitry connected to the RFB pin. Locally

bypass this pin to ground with a capacitor.

GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed

pad provides both electrical contact to ground and good

thermal contact to the printed circuit board. Solder the

exposed pad directly to the ground plane.

SW (Pin 5): Drain of the Internal DMOS Power Switch.

Minimize trace area at this pin to reduce EMI and voltage

spikes.

RFB (Pin 6): Input Pin for External Feedback Resistor.

Connect a resistor from this pin to the transformer primary

SW pin. The ratio of the RFB resistor to the RREF resistor,

times the internal voltage reference, determines the output

voltage (plus the effect of any non-unity transformer turns

ratio). Minimize trace area at this pin.

RREF (Pin 7): Input Pin for External Ground Referred Ref-

erence Resistor. The resistor at this pin should be in the

range of 10k, but for convenience in selecting a resistor

divider ratio, the value may range from 9.09k to 11.0k.

TC (Pin 8): Output Voltage Temperature Compensation. The

voltage at this pin is proportional to absolute temperature

(PTAT) with temperature coefficient equal to 3.35mV/°K,

i.e., equal to 1V at room temperature 25°C. The TC pin

voltage can be used to estimate the LT8302 junction tem-

perature. Connect a resistor from this pin to the RREF pin

to compensate the output diode temperature coefficient.

LT8302

8302fa

For more information www.linear.com/LT8302

block DiagraM

–

DRIVER

INTVCC

VIN

N:1

A2 RSENSE

8302 BD

RREF

REN2

REN1

RTC

RFB

–

1.214V

OSCILLATOR

LDO

BOUNDARY

DETECTOR

START-UP,

REFERENCE,

CONTROL

PTAT

VOLTAGE

R M1

GND

4, EXPOSED PAD PIN 9

VIN

CIN

RFB

L1A L1B COUT

DOUT

VOUT+

VOUT–

25µA

INTVCC

1EN/UVLO

CINTVCC

2.5µA

1:4

–

•

LT8302

8302fa

For more information www.linear.com/LT8302

operaTion

The LT8302 is a current mode switching regulator IC

designed specially for the isolated flyback topology. The

key problem in isolated topologies is how to communicate

the output voltage information from the isolated secondary

side of the transformer to the primary side for regulation.

Historically, opto-isolators or extra transformer windings

communicate this information across the isolation bound-

ary. Opto-isolator circuits waste output power, and the

extra components increase the cost and physical size of

the power supply. Opto-isolators can also cause system

issues due to limited dynamic response, nonlinearity, unit-

to-unit variation and aging over lifetime. Circuits employing

extra transformer windings also exhibit deficiencies, as

using an extra winding adds to the transformer’s physical

size and cost, and dynamic response is often mediocre.

The LT8302 samples the isolated output voltage through

the primary-side flyback pulse waveform. In this manner,

neither opto-isolator nor extra transformer winding is re-

quired for regulation. Since the LT8302 operates in either

boundary conduction mode or discontinuous conduction

mode, the output voltage is always sampled on the SW

pin when the secondary current is zero. This method im-

proves load regulation without the need of external load

compensation components.

The LT8302 is a simple to use micropower isolated fly-

back converter housed in a thermally enhanced 8-lead

SO package. The output voltage is programmed with two

external resistors. An optional TC resistor provides easy

output diode temperature compensation. By integrating

the loop compensation and soft-start inside, the part

reduces the number of external components. As shown

in the Block Diagram, many of the blocks are similar to

those found in traditional switching regulators including

reference, regulators, oscillator, logic, current amplifier,

current comparator, driver, and power switch. The novel

sections include a flyback pulse sense circuit, a sample-

and-hold error amplifier, and a boundary mode detector,

as well as the additional logic for boundary conduction

mode, discontinuous conduction mode, and low ripple

Burst Mode operation.

Quasi-Resonant Boundary Mode Operation

The LT8302 features quasi-resonant boundary conduction

mode operation at heavy load, where the chip turns on the

primary power switch when the secondary current is zero

and the SW rings to its valley. Boundary conduction mode

is a variable frequency, variable peak-current switching

scheme. The power switch turns on and the transformer

primary current increases until an internally controlled peak

current limit. After the power switch turns off, the voltage

on the SW pin rises to the output voltage multiplied by

the primary-to-secondary transformer turns ratio plus the

input voltage. When the secondary current through the

output diode falls to zero, the SW pin voltage collapses

and rings around VIN. A boundary mode detector senses

this event and turns the power switch back on at its valley.

LT8302

8302fa

For more information www.linear.com/LT8302

operaTion

Boundary conduction mode returns the secondary current

to zero every cycle, so parasitic resistive voltage drops

do not cause load regulation errors. Boundary conduc-

tion mode also allows the use of smaller transformers

compared to continuous conduction mode and does not

exhibit subharmonic oscillation.

Discontinuous Conduction Mode Operation

As the load gets lighter, boundary conduction mode in-

creases the switching frequency and decreases the switch

peak current at the same ratio. Running at a higher switching

frequency up to several MHz increases switching and gate

charge losses. To avoid this scenario, the LT8302 has an

additional internal oscillator, which clamps the maximum

switching frequency to be less than 380kHz. Once the

switching frequency hits the internal frequency clamp,

the part starts to delay the switch turn-on and operates

in discontinuous conduction mode.

Low Ripple Burst Mode Operation

Unlike traditional flyback converters, the LT8302 has to

turn on and off at least for a minimum amount of time

and with a minimum frequency to allow accurate sampling

of the output voltage. The inherent minimum switch cur-

rent limit and minimum switch-off time are necessary to

guarantee the correct operation of specific applications.

As the load gets very light, the LT8302 starts to fold back

the switching frequency while keeping the minimum switch

current limit. So the load current is able to decrease while

still allowing minimum switch-off time for the sample-and-

hold error amplifier. Meanwhile, the part switches between

sleep mode and active mode, thereby reducing the effec-

tive quiescent current to improve light load efficiency. In

this condition, the LT8302 runs in low ripple Burst Mode

operation. The typical 12kHz minimum switching frequency

determines how often the output voltage is sampled and

also the minimum load requirement.

LT8302

8302fa

For more information www.linear.com/LT8302

applicaTions inForMaTion

Output Voltage

The RFB and RREF resistors as depicted in the Block Diagram

are external resistors used to program the output voltage.

The LT8302 operates similar to traditional current mode

switchers, except in the use of a unique flyback pulse

sense circuit and a sample-and-hold error amplifier, which

sample and therefore regulate the isolated output voltage

from the flyback pulse.

Operation is as follows: when the power switch M1 turns

off, the SW pin voltage rises above the VIN supply. The

amplitude of the flyback pulse, i.e., the difference between

the SW pin voltage and VIN supply, is given as:

VFLBK = (VOUT + VF + ISEC • ESR) • NPS

VF = Output diode forward voltage

ISEC = Transformer secondary current

ESR = Total impedance of secondary circuit

NPS = Transformer effective primary-to-secondary

turns ratio

The flyback voltage is then converted to a current, IRFB,

by the RFB resistor and the flyback pulse sense circuit

(M2 and M3). This current, IRFB, also flows through the

RREF resistor to generate a ground-referred voltage. The

resulting voltage feeds to the inverting input of the sample-

and-hold error amplifier. Since the sample-and-hold error

amplifier samples the voltage when the secondary current

is zero, the (ISEC • ESR) term in the VFLBK equation can be

assumed to be zero.

The internal reference voltage, VREF, 1.00V, feeds to the

noninverting input of the sample-and-hold error ampli-

fier. The relatively high gain in the overall loop causes the

voltage at the RREF pin to be nearly equal to the internal

reference voltage VREF. The resulting relationship between

VFLBK and VREF can be expressed as:

VFLBK

RFB

⎛

⎝

⎜⎞

⎠

⎟•RREF =VREF or

VFLBK =VREF •RFB

RREF

⎛

⎝

⎜⎞

⎠

⎟

VREF = Internal reference voltage 1.00V

Combination with the previous VFLBK equation yields an

equation for VOUT, in terms of the RFB and RREF resistors,

transformer turns ratio, and diode forward voltage:

VOUT =VREF •RFB

RREF

⎛

⎝

⎜⎞

⎠

⎟•1

NPS

⎛

⎝

⎜⎞

⎠

⎟– VF

Output Temperature Compensation

The first term in the VOUT equation does not have tempera-

ture dependence, but the output diode forward voltage, VF,

has a significant negative temperature coefficient (–1mV/°C

to –2mV/°C). Such a negative temperature coefficient pro-

duces approximately 200mV to 300mV voltage variation

on the output voltage across temperature.

For higher voltage outputs, such as 12V and 24V, the

output diode temperature coefficient has a negligible ef-

fect on the output voltage regulation. For lower voltage

outputs, such as 3.3V and 5V, however, the output diode

temperature coefficient does count for an extra 2% to 5%

output voltage regulation.

The LT8302 junction temperature usually tracks the output

diode junction temperature to the first order. To compensate

the negative temperature coefficient of the output diode,

a resistor, RTC, connected between the TC and RREF pins

generates a proportional-to-absolute-temperature (PTAT)

current. The PTAT current is zero at 25°C, flows into the

RREF pin at hot temperature, and flows out of the RREF pin

at cold temperature. With the RTC resistor in place, the

output voltage equation is revised as follows:

VOUT =VREF •RFB

RREF

•1

NPS

– VFTO

( )

– VTC / T

( )

•

T –TO

( )

•RFB

RTC

•1

NPS

– VF/ T

( )

•T–TO

( )

TO=Room temperature 25°

VF/ T

( )

=Output diode forward voltage

temperature coefficient

/ T

( )

=3.35mV/ C

LT8302

8302fa

For more information www.linear.com/LT8302

applicaTions inForMaTion

To cancel the output diode temperature coefficient, the

following two equations should be satisfied:

VOUT =VREF •RFB

RREF

•1

NPS

– VFTO

( )

VTC/ T

( )

•RFB

RTC

•1

NPS

=– VF/ T

( )

Selecting Actual RREF, RFB, RTC Resistor Values

The LT8302 uses a unique sampling scheme to regulate

the isolated output voltage. Due to the sampling nature,

the scheme contains repeatable delays and error sources,

which will affect the output voltage and force a re-evaluation

of the RFB and RTC resistor values. Therefore, a simple

2-step sequential process is recommended for selecting

resistor values.

Rearrangement of the expression for VOUT in the previous

sections yields the starting value for RFB:

RFB =RREF •NPS •VOUT +VFTO

( )

REF

VOUT = Output voltage

VF (TO) = Output diode forward voltage at 25°C = ~0.3V

NPS = Transformer effective primary-to-secondary

turns ratio

The equation shows that the RFB resistor value is indepen-

dent of the RTC resistor value. Any RTC resistor connected

between the TC and RREF pins has no effect on the output

voltage setting at 25°C because the TC pin voltage is equal

to the RREF regulation voltage at 25°C.

The RREF resistor value should be approximately 10k

because the LT8302 is trimmed and specified using this

value. If the RREF resistor value varies considerably from

10k, additional errors will result. However, a variation in

RREF up to 10% is acceptable. This yields a bit of freedom

in selecting standard 1% resistor values to yield nominal

RFB/RREF ratios.

First, build and power up the application with the starting

RREF, RFB values (no RTC resistor yet) and other compo-

nents connected, and measure the regulated output volt-

age, VOUT(MEAS). The new RFB value can be adjusted to:

RFB(NEW) =

OUT

VOUT(MEAS)

•RFB

Second, with a new RFB resistor value selected, the output

diode temperature coefficient in the application can be

tested to determine the RTC value. Still without the RTC

resistor, the VOUT should be measured over temperature

at a desired target output load. It is very important for

this evaluation that uniform temperature be applied to

both the output diode and the LT8302. If freeze spray or

a heat gun is used, there can be a significant mismatch

in temperature between the two devices that causes sig-

nificant error. Attempting to extrapolate the data from a

diode data sheet is another option if there is no method

to apply uniform heating or cooling such as an oven. With

at least two data points spreading across the operating

temperature range, the output diode temperature coef-

ficient can be determined by:

–δVF/δT

( )

OUT

( )

– V

OUT

( )

T1– T2

Using the measured output diode temperature coefficient,

an exact RTC value can be selected with the following

equation:

RTC =δVTC/δT

( )

–δVF/δT

( )

•RFB

NPS

⎛

⎝

⎜⎞

⎠

⎟

Once the RREF, RFB, and RTC values are selected, the regula-

tion accuracy from board to board for a given application

will be very consistent, typically under ±5% when includ-

ing device variation of all the components in the system

(assuming resistor tolerances and transformer windings

matching within ±1%). However, if the transformer or

the output diode is changed, or the layout is dramatically

altered, there may be some change in VOUT.

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Output Power

A flyback converter has a complicated relationship between

the input and output currents compared to a buck or a

boost converter. A boost converter has a relatively constant

maximum input current regardless of input voltage and a

buck converter has a relatively constant maximum output

current regardless of input voltage. This is due to the

continuous non-switching behavior of the two currents. A

flyback converter has both discontinuous input and output

currents which make it similar to a nonisolated buck-boost

converter. The duty cycle will affect the input and output

currents, making it hard to predict output power. In ad-

dition, the winding ratio can be changed to multiply the

output current at the expense of a higher switch voltage.

The graphs in Figures 1 to 4 show the typical maximum

output power possible for the output voltages 3.3V, 5V,

12V, and 24V. The maximum output power curve is the

calculated output power if the switch voltage is 50V dur-

ing the switch-off time. 15V of margin is left for leakage

inductance voltage spike. To achieve this power level at

a given input, a winding ratio value must be calculated

to stress the switch to 50V, resulting in some odd ratio

values. The curves below the maximum output power

curve are examples of common winding ratio values and

the amount of output power at given input voltages.

One design example would be a 5V output converter with

a minimum input voltage of 8V and a maximum input volt-

age of 32V. A three-to-one winding ratio fits this design

example perfectly and outputs equal to 15.3W at 32V but

lowers to 7.7W at 8V.

Figure 1. Output Power for 3.3V Output

Figure 2. Output Power for 5V Output

Figure 3. Output Power for 12V Output

Figure 4. Output Power for 24V Output

INPUT VOLTAGE (V)

OUTPUT POWER (W)

8302 F02

010 20 30

MAXIMUM

OUTPUT POWER

N = 3:1

N = 1:1

N = 4:1

N = 2:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

8302 F03

010 20 30

N = 1:1

MAXIMUM

OUTPUT POWER

N = 3:2

N = 1:2

N = 2:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

8302 F04

010 20 30

N = 1:2

MAXIMUM

OUTPUT POWER

N = 2:3

N = 1:3

N = 1:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

8302 F01

010 20 30

N = 6:1

MAXIMUM

OUTPUT POWER

N = 4:1

N = 2:1

N = 3:1

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The equations below calculate output power:

POUT = η • VIN • D • ISW(MAX) • 0.5

η = Efficiency = ~85%

D=Duty Cycle =VOUT +VF

( )

•NPS

VOUT +VF

( )

•NPS +VIN

ISW(MAX) = Maximum switch current limit = 3.6A (MIN)

Primary Inductance Requirement

The LT8302 obtains output voltage information from the

reflected output voltage on the SW pin. The conduction

of secondary current reflects the output voltage on the

primary SW pin. The sample-and-hold error amplifier needs

a minimum 350ns to settle and sample the reflected output

voltage. In order to ensure proper sampling, the second-

ary winding needs to conduct current for a minimum of

350ns. The following equation gives the minimum value

for primary-side magnetizing inductance:

LPRI ≥tOFF(MIN) •NPS •VOUT +VF

( )

ISW(MIN)

tOFF(MIN) = Minimum switch-off time = 350ns (TYP)

ISW(MIN) = Minimum switch current limit = 0.87A (TYP)

In addition to the primary inductance requirement for

the minimum switch-off time, the LT8302 has minimum

switch-on time that prevents the chip from turning on

the power switch shorter than approximately 160ns. This

minimum switch-on time is mainly for leading-edge blank-

ing the initial switch turn-on current spike. If the inductor

current exceeds the desired current limit during that time,

oscillation may occur at the output as the current control

loop will lose its ability to regulate. Therefore, the following

equation relating to maximum input voltage must also be

followed in selecting primary-side magnetizing inductance:

LPRI ≥

ON(MIN)

•V

IN(MAX)

ISW(MIN)

tON(MIN) = Minimum switch-on time = 160ns (TYP)

In general, choose a transformer with its primary mag-

netizing inductance about 40% to 60% larger than the

minimum values calculated above. A transformer with

much larger inductance will have a bigger physical size

and may cause instability at light load.

Selecting a Transformer

Transformer specification and design is perhaps the most

critical part of successfully applying the LT8302. In addition

to the usual list of guidelines dealing with high frequency

isolated power supply transformer design, the following

information should be carefully considered.

Linear Technology has worked with several leading mag-

netic component manufacturers to produce pre-designed

flyback transformers for use with the LT8302. Table 1

shows the details of these transformers.

Table 1. Predesigned Transformers–Typical Specifications

TRANSFORMER

PART NUMBER

DIMENSIONS

(W × L × H) (mm)

LPRI

(µH)

LLKG

(µH) NP:NS

RPRI

(mΩ)

RSEC

(mΩ) VENDOR

TARGET APPLICATION

VIN (V) VOUT (V) IOUT (A)

750311625 17.75 × 13.46 × 12.70 9 0.35 4:1 43 6 Würth Elektronik 8 to 32 3.3 2.1

750311564 17.75 × 13.46 × 12.70 9 0.12 3:1 36 7 Würth Elektronik 8 to 32 5 1.5

750313441 15.24 × 13.34 x 11.43 9 0.6 2:1 75 18 Würth Elektronik 8 to 32 5 1.3

750311624 17.75 × 13.46 × 12.70 9 0.18 3:2 34 21 Würth Elektronik 8 to 32 8 0.9

750313443 15.24 × 13.34 × 11.43 9 0.3 1:1:1 85 100 Würth Elektronik 8 to 36 ±12 0.3

750313445 15.24 × 13.34 × 11.43 9 0.25 1:2 85 190 Würth Elektronik 8 to 36 24 0.3

750313457 15.24 × 13.34 × 11.43 9 0.25 1:4 85 770 Würth Elektronik 8 to 36 48 0.15

750313460 15.24 × 13.34 × 11.43 12 0.7 4:1 85 11 Würth Elektronik 4 to 18 5 0.9

750311342 15.24 × 13.34 × 11.43 15 0.44 2:1 85 22 Würth Elektronik 4 to 18 12 0.4

750313439 15.24 × 13.34 × 11.43 12 0.6 2:1 115 28 Würth Elektronik 18 to 42 3.3 2.1

750313442 15.24 × 13.34 × 11.43 12 0.75 3:2 150 53 Würth Elektronik 18 to 42 5 1.6

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Turns Ratio

Note that when choosing an RFB/RREF resistor ratio to set

output voltage, the user has relative freedom in selecting

a transformer turns ratio to suit a given application. In

contrast, the use of simple ratios of small integers, e.g.,

3:1, 2:1, 1:1, etc., provides more freedom in settling total

turns and mutual inductance.

Typically, choose the transformer turns ratio to maximize

available output power. For low output voltages (3.3V

or 5V), a N:1 turns ratio can be used with multiple pri-

mary windings relative to the secondary to maximize the

transformer’s current gain (and output power). However,

remember that the SW pin sees a voltage that is equal

to the maximum input supply voltage plus the output

voltage multiplied by the turns ratio. In addition, leakage

inductance will cause a voltage spike (VLEAKAGE) on top of

this reflected voltage. This total quantity needs to remain

below the 65V absolute maximum rating of the SW pin to

prevent breakdown of the internal power switch. Together

these conditions place an upper limit on the turns ratio,

NPS, for a given application. Choose a turns ratio low

enough to ensure

NPS <

65V – V

IN(MAX)

– V

LEAKAGE

OUT

For larger N:1 values, choose a transformer with a larger

physical size to deliver additional current. In addition,

choose a large enough inductance value to ensure that

the switch-off time is long enough to accurately sample

the output voltage.

For lower output power levels, choose a 1:1 or 1:N trans-

former for the absolute smallest transformer size. A 1:N

transformer will minimize the magnetizing inductance

(and minimize size), but will also limit the available output

power. A higher 1:N turns ratio makes it possible to have

very high output voltages without exceeding the breakdown

voltage of the internal power switch.

The turns ratio is an important element in the isolated

feedback scheme, and directly affects the output voltage

accuracy. Make sure the transformer manufacturer speci-

fies turns ratio accuracy within ±1%.

Saturation Current

The current in the transformer windings should not exceed

its rated saturation current. Energy injected once the core is

saturated will not be transferred to the secondary and will

instead be dissipated in the core. When designing custom

transformers to be used with the LT8302, the saturation

current should always be specified by the transformer

manufacturers.

Winding Resistance

Resistance in either the primary or secondary windings

will reduce overall power efficiency. Good output voltage

regulation will be maintained independent of winding re-

sistance due to the boundary/discontinuous conduction

mode operation of the LT8302.

Leakage Inductance and Snubbers

ransformer leakage inductance on either the primary or

secondary causes a voltage spike to appear on the primary

after the power switch turns off. This spike is increasingly

prominent at higher load currents where more stored en-

ergy must be dissipated. It is very important to minimize

transformer leakage inductance.

When designing an application, adequate margin should

be kept for the worst-case leakage voltage spikes even

under overload conditions. In most cases shown in Fig-

ure5, the reflected output voltage on the primary plus VIN

should be kept below 50V. This leaves at least 15V margin

for the leakage spike across line and load conditions. A

larger voltage margin will be required for poorly wound

transformers or for excessive leakage inductance.

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tOFF > 350ns

VLEAKAGE

VSW

<65V

<50V

TIME 8302 F05

tSP < 250ns

Figure 5. Maximum Voltages for SW Pin Flyback Waveform

In addition to the voltage spikes, the leakage inductance

also causes the SW pin ringing for a while after the power

switch turns off. To prevent the voltage ringing falsely trig-

ger boundary mode detector, the LT8302 internally blanks

the boundary mode detector for approximately 250ns.

Any remaining voltage ringing after 250ns may turn the

power switch back on again before the secondary current

falls to zero. In this case, the LT8302 enters continuous

conduction mode. So the leakage inductance spike ringing

should be limited to less than 250ns.

To clamp and damp the leakage voltage spikes, a

(RC + DZ) snubber circuit in Figure6 is recommended.

The RC (resistor-capacitor) snubber quickly damps the

voltage spike ringing and provides great load regulation

and EMI performance. And the DZ (diode-Zener) ensures

well defined and consistent clamping voltage to protect

SW pin from exceeding its 65V absolute maximum rating.

Figure 6. (RC + DZ) Snubber Circuit

8302 F06

Lℓ

•

then add capacitance until the period of the ringing is 1.5

to 2 times longer. The change in period determines the

value of the parasitic capacitance, from which the para-

sitic inductance can be also determined from the initial

period. Once the value of the SW node capacitance and

inductance is known, a series resistor can be added to

the snubber capacitance to dissipate power and critically

damp the ringing. The equation for deriving the optimal

series resistance using the observed periods ( tPERIOD and

tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is:

CPAR =

SNUBBER

tPERIOD(SNUBBED)

tPERIOD

⎛

⎝

⎜⎞

⎠

⎟

–

LPAR =tPERIOD2

CPAR •4π2

RSNUBBER =LPAR

CPAR

Note that energy absorbed by the RC snubber will be

converted to heat and will not be delivered to the load.

In high voltage or high current applications, the snubber

needs to be sized for thermal dissipation. A 470pF capaci-

tor in series with a 39Ω resistor is a good starting point.

For the DZ snubber, proper care should be taken when

choosing both the diode and the Zener diode. Schottky

diodes are typically the best choice, but some PN diodes

can be used if they turn on fast enough to limit the leak-

age inductance spike. Choose a diode that has a reverse-

voltage rating higher than the maximum SW pin voltage.

The Zener diode breakdown voltage should be chosen to

balance power loss and switch voltage protection. The best

compromise is to choose the largest voltage breakdown

with 5V margin. Use the following equation to make the

proper choice:

VZENNER(MAX) ≤ 60V – VIN(MAX)

For an application with a maximum input voltage of 32V,

choose a 24V Zener diode, the VZENER(MAX) of which is

around 26V and below the 28V maximum. The power loss

in the DZ snubber determines the power rating of the Zener

diode. A 1.5W Zener diode is typically recommended.

The recommended approach for designing an RC snub-

ber is to measure the period of the ringing on the SW pin

when the power switch turns off without the snubber and

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Undervoltage Lockout (UVLO)

A resistive divider from VIN to the EN/UVLO pin imple-

ments undervoltage lockout (UVLO). The EN/UVLO enable

falling threshold is set at 1.214V with 14mV hysteresis. In

addition, the EN/UVLO pin sinks 2.5µA when the voltage

on the pin is below 1.214V. This current provides user

programmable hysteresis based on the value of R1. The

programmable UVLO thresholds are:

VIN(UVLO+)=1.228V •R1+R2

( )

R2 +2.5µA •R

VIN(UVLO–)=1.214V •R1+R2

( )

Figure 7 shows the implementation of external shutdown

control while still using the UVLO function. The NMOS

grounds the EN/UVLO pin when turned on, and puts the

LT8302 in shutdown with quiescent current less than 2µA.

LT8302

GND

EN/UVLO

RUN/STOP

CONTROL

(OPTIONAL)

VIN

8302 F07

Figure 7. Undervoltage Lockout (UVLO)

Minimum Load Requirement

The LT8302 samples the isolated output voltage from

the primary-side flyback pulse waveform. The flyback

pulse occurs once the primary switch turns off and the

secondary winding conducts current. In order to sample

the output voltage, the LT8302 has to turn on and off for a

minimum amount of time and with a minimum frequency.

The LT8302 delivers a minimum amount of energy even

during light load conditions to ensure accurate output volt-

age information. The minimum energy delivery creates a

minimum load requirement, which can be approximately

estimated as:

ILOAD(MIN) =

•I

SW(MIN)

•f

MIN

2•VOUT

LPRI = Transformer primary inductance

ISW(MIN) = Minimum switch current limit = 0.96A (MAX)

fMIN = Minimum switching frequency = 12.7kHz (MAX)

The LT8302 typically needs less than 0.5% of its full output

power as minimum load. Alternatively, a Zener diode with

its breakdown of 10% higher than the output voltage can

serve as a minimum load if pre-loading is not acceptable.

For a 5V output, use a 5.6V Zener with cathode connected

to the output.

Output Short Protection

When the output is heavily overloaded or shorted to ground,

the reflected SW pin waveform rings longer than the in-

ternal blanking time. After the 350ns minimum switch-off

time, the excessive ringing falsely triggers the boundary

mode detector and turns the power switch back on again

before the secondary current falls to zero. Under this

condition, the LT8302 runs into continuous conduction

mode at 380kHz maximum switching frequency. If the

sampled RREF voltage is still less than 0.6V after 11ms

(typ) soft-start timer, the LT8302 initiates a new soft-start

cycle. If the sampled RREF voltage is larger than 0.6V after

11ms, the switch current may run away and exceed the

4.5A maximum current limit. Once the switch current hits

7.2A over current limit, the LT8302 also initiates a new

soft-start cycle. Under either condition, the new soft-start

cycle throttles back both the switch current limit and switch

frequency. The output short-circuit protection prevents the

switch current from running away and limits the average

output diode current.

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Design Example

Use the following design example as a guide to designing

applications for the LT8302. The design example involves

designing a 5V output with a 1.5A load current and an

input range from 8V to 32V.

VIN(MIN) = 8V, VIN(NOM) = 12V, VIN(MAX) = 32V,

VOUT = 5V, IOUT = 1.5A

Step 1: Select the transformer turns ratio.

NPS <

65V – V

IN(MAX)

– V

LEAKAGE

VOUT +V

VLEAKAGE = Margin for transformer leakage spike = 15V

VF = Output diode forward voltage = ~0.3V

Example:

NPS <

65V – 32V – 15V

5V +0.3V

=3.4

The choice of transformer turns ratio is critical in determin-

ing output current capability of the converter. Table2 shows

the switch voltage stress and output current capability at

different transformer turns ratio.

Table 2. Switch Voltage Stress and Output Current Capability vs

Turns Ratio

NPS

VSW(MAX) at

VIN(MAX) (V)

IOUT(MAX) at

VIN(MIN) (A) DUTY CYCLE (%)

1:1 37.3 0.92 14-40

2:1 42.6 1.31 25-57

3:1 47.9 1.53 33-67

Clearly, only NPS = 3 can meet the 1.5A output current

requirement, so NPS = 3 is chosen as the turns ratio in

this example.

Step 2: Determine the primary inductance.

Primary inductance for the transformer must be set above

a minimum value to satisfy the minimum switch-off and

switch-on time requirements:

LPRI ≥tOFF(MIN) •NPS •VOUT +V

( )

ISW(MIN)

LPRI ≥tON(MIN) •V

IN(MAX)

ISW(MIN)

tOFF(MIN) = 350ns

tON(MIN) = 160ns

ISW(MIN) = 0.87A

Example:

LPRI ≥350ns •3•5V +0.3V

( )

0.87A =6.4µH

LPRI ≥160ns •32V

0.87A

=5.9µH

Most transformers specify primary inductance with a toler-

ance of ±20%. With other component tolerance considered,

choose a transformer with its primary inductance 40% to

60% larger than the minimum values calculated above.

LPRI = 9µH is then chosen in this example.

Once the primary inductance has been determined, the

maximum load switching frequency can be calculated as:

fSW =

tON +tOFF

LPRI •ISW

+LPRI •ISW

NPS •VOUT +VF

( )

ISW =VOUT •IOUT •2

η•V

•D

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Example:

5V +0.3V

( )

•3

5V +0.3V

( )

•3+12V =0.57

ISW =5V •1.5A •2

0.8 •12V •0.57

=277kHz

The transformer also needs to be rated for the correct

saturation current level across line and load conditions.

A saturation current rating larger than 7A is necessary

to work with the LT8302. The 750311564 from Würth is

chosen as the flyback transformer.

Step 3: Choose the output diode.

Tw o main criteria for choosing the output diode include

forward current rating and reverse-voltage rating. The

maximum load requirement is a good first-order guess

at the average current requirement for the output diode.

Under output short-circuit condition, the output diode

needs to conduct much higher current. Therefore, a con-

servative metric is 60% of the maximum switch current

limit multiplied by the turns ratio:

IDIODE(MAX) = 0.6 • ISW(MAX) • NPS

Example:

IDIODE(MAX) = 8.1A

Next calculate reverse voltage requirement using maxi-

mum VIN:

REVERSE =VOUT +

IN(MAX)

Example:

REVERSE =5V +

32V

=15.7V

The PDS835L (8A, 35V diode) from Diodes Inc. is chosen.

Step 4: Choose the output capacitor.

The output capacitor should be chosen to minimize the

output voltage ripple while considering the increase in size

and cost of a larger capacitor. Use the following equation

to calculate the output capacitance:

COUT =

PRI

•I

2•VOUT •ΔVOUT

Example:

Design for output voltage ripple less than ±1% of VOUT,

i.e., 100mV.

COUT =9µH •4.5A

( )

2•5V •0.1V

=182µF

Remember ceramic capacitors lose capacitance with ap-

plied voltage. The capacitance can drop to 40% of quoted

capacitance at the maximum voltage rating. So a 220µF,

6.3V rating ceramic capacitor is chosen.

Step 5: Design snubber circuit.

The snubber circuit protects the power switch from leak-

age inductance voltage spike. A (RC + DZ) snubber is

recommended for this application. A 470pF capacitor in

series with a 39Ω resistor is chosen as the RC snubber.

The maximum Zener breakdown voltage is set according

to the maximum VIN:

VZENNER(MAX) ≤ 60V – VIN(MAX)

Example:

VZENNER(MAX) ≤ 60V – 32V = 28V

A 24V Zener with a maximum of 26V will provide optimal

protection and minimize power loss. So a 24V, 1.5W Zener

from Central Semiconductor (CMZ5934B) is chosen.

Choose a diode that is fast and has sufficient reverse

voltage breakdown:

VREVERSE > VSW(MAX)

VSW(MAX) = VIN(MAX) + VZENNER(MAX)

Example:

VREVERSE > 60V

A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen.

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Step 6: Select the RREF and RFB resistors.

Use the following equation to calculate the starting values

for RREF and RFB:

RFB =RREF •NPS •VOUT +V

FTO

( )

REF

=10k

Example:

RFB =

10k •3•5V +0.3V

( )

1.00V

=159k

For 1% standard values, a 158k resistor is chosen.

Step 7: Adjust RFB resistor based on output voltage.

Build and power up the application with application com-

ponents and measure the regulated output voltage. Adjust

RFB resistor based on the measured output voltage:

RFB(NEW) =

OUT

VOUT(MEASURED)

•RFB

Example:

RFB =

5.14V

•158k =154k

Step 8: Select RTC resistor based on output voltage

temperature variation.

Measure output voltage in a controlled temperature envi-

ronment like an oven to determine the output temperature

coefficient. Measure output voltage at a consistent load

current and input voltage, across the operating tempera-

ture range.

Calculate the temperature coefficient of VF:

–δV

F/δT

( )

OUT

( )

– V

OUT

( )

T1– T2

RTC =3.35mV/°C

–δV

F/δT

( )

•RFB

NPS

⎛

⎝

⎜⎞

⎠

⎟

Example:

–δV

F/δT

( )

5.189V – 5.041V

100°C – 0°C

( )

=1.48mV / °C

RTC =3.35mV/°C

1.48mV/°C•154

⎛

⎝

⎜⎞

⎠

⎟=115k

Step 9: Select the EN/UVLO resistors.

Determine the amount of hysteresis required and calculate

R1 resistor value:

VIN(HYS) = 2.5µA • R1

Example:

Choose 2V of hysteresis, R1 = 806k

Determine the UVLO thresholds and calculate R2 resistor

value:

IN(UVLO+)=

1.228V •R1+R2

( )

+2.5µA •R

Example:

Set VIN UVLO rising threshold to 7.5V:

R2 = 232k

VIN(UVLO+) = 7.5V

VIN(UNLO–) = 5.5V

Step 10: Ensure minimum load.

The theoretical minimum load can be approximately

estimated as:

ILOAD(MIN) =9µH •0.96A

( )

2•12.7kHz

2•5V

=10.5mA

Remember to check the minimum load requirement in

real application. The minimum load occurs at the point

where the output voltage begins to climb up as the con-

verter delivers more energy than what is consumed at

the output. The real minimum load for this application is

about 10mA. In this example, a 500Ω resistor is selected

as the minimum load.

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Typical applicaTions

8V to 32VIN/12VOUT Isolated Flyback Converter

8V to 32VIN/3.3VOUT Isolated Flyback Converter

Efficiency vs Load Current Load and Line Regulation

AMBIENT TEMPERATURE (°C)

–50

OUTPUT VOLTAGE (V)

3.30

3.40

150

8302 TA03b

3.20

3.10 050 100

–25 25 75 125

3.50

3.25

3.35

3.15

3.45

RTC = 105k

RTC = OPEN

VIN = 12V

IOUT = 1A

Output Temperature Variation

LOAD CURRENT (mA)

EFFICIENCY (%)

200 400 600 800 1000

8302 TA02b

1200

VIN = 12V

VIN = 24V

LOAD CURRENT (mA)

11.2

OUTPUT VOLTAGE (V)

11.4

11.6

11.8

12.0

12.2

12.4

200 400 600 800 1000 1200

8302 TA02c

VIN = 12V

VIN = 24V

VIN

LT8302

9µH

VIN

8V TO 32V

1:1

9µH

RFB

RREF

470pF

10µF

1µF

47µF

VOUT+

12V

5mA TO 0.8A (VIN = 12V)

5mA TO 1.1A (VIN = 24V)

VOUT–

39Ω

121k

•

OPEN

232k

806k

10k

D1: DIODES DFLS1100

D2: DIODES PDS540

T1: WURTH 750313443

Z1: CENTRAL CMZ5934B

8302 TA02a

EN/UVLO

GND

INTVCC

VIN

LT8302

9µH

VIN

8V TO 32V

4:1

0.56µH

RFB

RREF

470pF

10µF

1µF

470µF

VOUT+

3.3V

20mA TO 2.7A (VIN = 12V)

20mA TO 3.8A (VIN = 24V)

VOUT–

39Ω

140k

•

105k

232k

806k

10k

D1: DIODES DFLS1100

D2: DIODES PDS1040L

T1: WURTH 750311625

Z1: CENTRAL CMZ5934B

8302 TA03

EN/UVLO

GND

INTVCC

LT8302

8302fa

For more information www.linear.com/LT8302

Typical applicaTions

8V to 36VIN/±12VOUT Isolated Flyback Converter

8V to 36VIN/24VOUT Isolated Flyback Converter

8V to 36VIN/48VOUT Isolated Flyback Converter

VIN

LT8302

9µH

VIN

8V TO 36V

1:1:1

9µH

RFB

RREF

470pF

10µF

1µF

22µF

VOUT1+

12V

5mA TO 0.4A (VIN = 12V)

5mA TO 0.55A (VIN = 24V)

VOUT2–

39Ω

121k

•

OPEN

232k

806k

10k

D1: DIODES DFLS1100

D2, D3: DIODES PDS360

T1: WURTH 750313443

Z1: CENTRAL CMZ5934B

8302 TA04

9µH

22µF

VOUT2+

12V

5mA TO 0.4A (VIN = 12V)

5mA TO 0.55A (VIN = 24V)

VOUT2–

•

EN/UVLO

GND

INTVCC

VIN

LT8302

9µH

VIN

8V TO 36V

1:2

36µH

RFB

RREF

470pF

10µF

1µF

10µF

VOUT+

24V

2.5mA TO 0.4A (VIN = 12V)

2.5mA TO 0.55A (VIN = 24V)

VOUT–

39Ω

121k

•

OPEN

232k

806k

10k

D1: DIODES DFLS1100

D2: DIODES SBR2U150SA

T1: WURTH 750313445

Z1: CENTRAL CMZ5934B

8302 TA05

EN/UVLO

GND

INTVCC

VIN

LT8302

9µH

VIN

8V TO 36V

1:4

144µH

RFB

RREF

470pF

10µF

1µF

2.2µF

VOUT+

48V

1.2mA TO 0.2A (VIN = 12V)

1.2mA TO 0.27A (VIN = 24V)

VOUT–

39Ω

121k

•

OPEN

232k

806k

10k

D1: DIODES DFLS1100

D2: DIODES SBR1U200P1

T1: WURTH 750313457

Z1: CENTRAL CMZ5934B

8302 TA06

EN/UVLO

GND

INTVCC

LT8302

8302fa

For more information www.linear.com/LT8302

Typical applicaTions

VIN

LT8302

9µH

VIN

8V TO 32V

3:1

1µH

RFB

RREF

470pF

D1 D2

10µF

1µF

220µF

10µF

4.7µF

VOUT+

5V/1.1A (VIN = 5V)

5V/2.0A (VIN = 12V)

5V/2.9A (VIN = 24V)

VOUT–

39Ω R7

5Ω

VCC

DRAIN

LT8309

GATE INTVCC

GND

154k

•

OPEN

232k

806k

2.1k

10k

D1: DIODES DFLS1100

D2: CENTRAL CMMSH1-60

M1: INFINEON BSC059N04LS

T1: WURTH 750311564

Z1: CENTRAL CMZ5934B

8302 TA07

EN/UVLO

GND

INTVCC

LOAD CURRENT (A)

EFFICIENCY (%)

1.5 2.5

8302 TA07b

0.5 1.0 2.0 3.0

8V to 32VIN/5VOUT Isolated Flyback Converter with LT8309

–4V to –42VIN/12VOUT Buck-Boost Converter

–18V to –42VIN/–12VOUT Negative Buck Converter

Efficiency vs Load Current

VIN SW

LT8302

12µH D1

GND

RFB

RREF

EN/UVLO

47µF

D1: DIODES PMEG6030EP

L1: WÜRTH 744770112

Z1: CENTRAL CMHZ5243B

1µF

VIN

–4V TO –42V

10µF

10k

8302 TA08a

VOUT

12V/0.45A (VIN = –5V)

12V/0.8A (VIN = –12V)

12V/1.1A (VIN = –24V)

12V/1.3A (VIN = –42V)

118k

INTVCC

LOAD CURRENT (mA)

EFFICIENCY (%)

200 400 800600 1000 1200 1400

8302 TA08b

VIN = –5V

VIN = –12V

VIN = –24V

VIN = –42V

VIN

LT8302

12µH

VOUT

–12V

1.8A

D1 Z1

RREF

EN/UVLO

RFB

EN/UVLO D1: DIODES PMEG6030EP

L1: WÜRTH 744770112

Z1: CENTRAL CMHZ5243B

1µF

VIN

–18V TO –42V

10µF

47µF

10k

8302 TA09a

118k

232k

806k

INTVCC

LOAD CURRENT (mA)

EFFICIENCY (%)

100

500 1000 1500 2000

8302 TA09b

VIN = –18V

VIN = –24V

VIN = –42V

LT8302

8302fa

For more information www.linear.com/LT8302

package DescripTion

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

.016 – .050

(0.406 – 1.270)

.010 – .020

(0.254 – 0.508)× 45°

0°– 8° TYP

.008 – .010

(0.203 – 0.254)

S8E 1013 REV A

.053 – .069

(1.346 – 1.752)

.014 – .019

(0.355 – 0.483)

TYP

.004 – .010

(0.101 – 0.254)

.080 – .098

(2.032 – 2.489)

.118 – .138

(2.997 – 3.505)

.050

(1.270)

BSC

1234

.150 – .157

(3.810 – 3.988)

NOTE 3

.005 (0.13) MAX

.189 – .197

(4.801 – 5.004)

NOTE 3

.228 – .244

(5.791 – 6.197)

.160 ±.005

(4.06 ±0.127)

.118

(2.99)

REF

RECOMMENDED SOLDER PAD LAYOUT

.045 ±.005

(1.143 ±0.127)

.050

(1.27)

BSC

INCHES

(MILLIMETERS)

NOTE:

1. DIMENSIONS IN

2. DRAWING NOT TO SCALE

3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)

S8E Package

8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad

(Reference LTC DWG # 05-08-1857 Rev A)

.089

(2.26)

REF

.030 ±.005

(0.76 ±0.127)

TYP

.245

(6.22)

MIN

LT8302

8302fa

For more information www.linear.com/LT8302

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

A 11/14 Modified IQ and IHYS Conditions

Modified LPRI Equation

Modified Schematic

Updated Related Parts

LT8302

8302fa

For more information www.linear.com/LT8302

LINEAR TECHNOLOGY CORPORATION 2013

LT 1114 REV A • PRINTED IN USA

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8302

relaTeD parTs

Typical applicaTion

4V to 42VIN/48VOUT Boost Converter

Efficiency vs Load Current

VIN SW

LT8302

22µH D1

GND

RFB

RREF

EN/UVLO C3

10µF

D1: DIODES PDS560

L1: WÜRTH 7443551221

Z1: CENTRAL CMHZ5262B

1µF

VIN

4V TO 42V

10µF

10k

8302 TA10a

VOUT

48V/1.4A (VIN = 42V)

48V/0.8A (VIN = 24V)

48V/0.4A (VIN = 12V)

48V/0.15A (VIN = 5V)

464k

INTVCC

LOAD CURRENT (mA)

EFFICIENCY (%)

100

250 500

8302 TA10b

1500750 1000 1250

VIN = 5V

VIN = 12V

VIN = 24V

VIN = 42V

PART NUMBER DESCRIPTION COMMENTS

LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A

Switch Low IQ Monolithic No-Opto Flyback 5-Lead TSOT-23

LT8300 100VIN Micropower Isolated Flyback Converter with

150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23

LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23

LT3573/LT3574

LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A

Switch

LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA

Switch, MSOP-16(12)

LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12)

LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components

LT3757A/LT3759

LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive

LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch