LT1713/LT1714 Single/Dual, 7ns, Low Power, 3V/5V/5V Rail-to-Rail Comparators DESCRIPTIO U FEATURES The LT(R)1713/LT1714 are UltraFastTM 7ns, single/dual comparators featuring rail-to-rail inputs, rail-to-rail complementary outputs and an output latch. Optimized for 3V and 5V power supplies, they operate over a single supply voltage range from 2.4V to 12V or from 2.4V to 6V dual supplies. Ultrafast: 7ns at 20mV Overdrive 8.5ns at 5mV Overdrive Rail-to-Rail Inputs Rail-to-Rail Complementary Outputs (TTL/CMOS Compatible) Specified at 2.7V, 5V and 5V Supplies Low Power (Per Comparator): 5mA Output Latch Inputs Can Exceed Supplies Without Phase Reversal LT1713: 8-Lead MSOP Package LT1714: 16-Lead Narrow SSOP Package The LT1713/LT1714 are designed for ease of use in a variety of systems. In addition to wide supply voltage flexibility, rail-to-rail input common mode range extends 100mV beyond both supply rails and the outputs are protected against phase reversal for inputs extending further beyond the rails. Also, the rail-to-rail inputs may be taken to opposite rails with no significant increase in input current. The rail-to-rail matched complementary outputs interface directly to TTL or CMOS logic and can sink 10mA to within 0.5V of GND or source 10mA to within 0.7V of V +. U APPLICATIO S High Speed Automatic Test Equipment Current Sense for Switching Regulators Crystal Oscillator Circuits High Speed Sampling Circuits High Speed A/D Converters Pulse Width Modulators Window Comparators Extended Range V/F Converters Fast Pulse Height/Width Discriminators Line Receivers High Speed Triggers The LT1713/LT1714 have internal TTL/CMOS compatible latches for retaining data at the outputs. Each latch holds data as long as its latch pin is held high. Latch pin hysteresis provides protection against slow moving or noisy latch signals. The LT1713 is available in the 8-lead MSOP package. The LT1714 is available in the 16-lead narrow SSOP package. , LTC and LT are registered trademarks of Linear Technology Corporation. UltraFast is a trademark of Linear Technology Corporation. U TYPICAL APPLICATIO A 4x NTSC Subcarrier Voltage-Tunable Crystal Oscillator 5V 1N4148 1M 47k* LT1004-2.5 1M* 3.9k* LT1713/LT1714 Propagation Delay vs Input Overdrive VIN 0V TO 5V 9.0 1k* 5V 1M 2k 390 MV-209 VARACTOR DIODE 1M 100pF PROPAGATION DELAY (ns) 8.5 0.047F C SELECT (CHOOSE FOR CORRECT PLL LOOP RESPONSE) Y1** 15pF 100pF + LT1713 - FREQUENCY OUTPUT 2k 200pF 171314 TA01 * 1% FILM RESISTOR ** NORTHERN ENGINEERING LABS C-2350N-14.31818MHz 8.0 tPD+ 7.5 TA = 25C V + = 5V V - = 0V VSTEP = 100mV tPD- 7.0 6.5 6.0 5.5 5.0 0 10 20 40 50 30 INPUT OVERDRIVE (mV) 60 171314 TA02 1 LT1713/LT1714 W W W AXI U U ABSOLUTE RATI GS (Note 1) Supply Voltage V + to V - ............................................................ 12.6V V + to GND ........................................................ 12.6V V - to GND .............................................- 10V to 0.3V Differential Input Voltage ................................... 12.6V Latch Pin Voltage ...................................................... 7V Input and Latch Current ..................................... 10mA Output Current (Continuous) .............................. 20mA Operating Temperature Range ................ - 40C to 85C Specified Temperature Range (Note 2) ... - 40C to 85C Junction Temperature .......................................... 150C Storage Temperature Range ................. - 65C to 150C Lead Temperature (Soldering, 10 sec).................. 300C U U W PACKAGE/ORDER I FOR ATIO ORDER PART NUMBER TOP VIEW V+ +IN -IN V- 1 2 3 4 8 7 6 5 Q OUT Q OUT GND LATCH ENABLE LT1713CMS8 LT1713IMS8 -IN A 1 +IN A 2 LATCH ENABLE A 15 GND V- 3 14 Q A V+ 4 13 Q A V+ 5 12 Q B - 6 11 Q B +IN B 7 -IN B 8 10 GND LATCH 9 ENABLE B V MS8 PACKAGE 8-LEAD PLASTIC MSOP MS8 PART MARKING TJMAX = 150C, JA = 250C/ W LTRD LTUK ORDER PART NUMBER TOP VIEW 16 LT1714CGN LT1714IGN GN PART MARKING 1714 1714I GN PACKAGE 16-LEAD PLASTIC SSOP TJMAX = 150C, JA = 120C/ W Consult factory for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. V + = 2.7V or V + = 5V, V - = 0V, VCM = V +/2, VLATCH = 0.8V, CLOAD = 10pF, VOVERDRIVE = 20mV, unless otherwise specified. SYMBOL PARAMETER V+ VOS CONDITIONS Positive Supply Voltage Range Input Offset Voltage (Note 4) VOS/T Input Offset Voltage Drift IOS Input Offset Current MIN RS = 50, VCM = V +/2 RS = 50, VCM = V +/2 (Note 11) RS = 50, VCM = 0V RS = 50, VCM = V + TYP 2.4 0.5 Input Bias Current (Note 5) VCM Input Voltage Range (Note 9) CMRR Common Mode Rejection Ratio 2 V + = 5V, 0V VCM 5V V + = 5V, 0V VCM 5V V + = 2.7V, 0V VCM 2.7V V + = 2.7V, 0V VCM 2.7V V 4 5 mV mV mV mV -7 - 15 - 0.1 V/C 5 60 58 57 55 UNITS 7 0.7 1 0.1 1 2 A A - 1.5 2 5 A A IB MAX V + + 0.1 70 70 V dB dB dB dB LT1713/LT1714 ELECTRICAL CHARACTERISTICS The denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. V + = 2.7V or V + = 5V, V - = 0V, VCM = V +/2, VLATCH = 0.8V, CLOAD = 10pF, VOVERDRIVE = 20mV, unless otherwise specified. SYMBOL PARAMETER CONDITIONS PSRR+ 2.4V V + 7V, VCM = 0V PSRR- AV Positive Power Supply Rejection Ratio Negative Power Supply Rejection Ratio MIN TYP 65 60 80 dB dB 65 60 80 dB dB 3 V/mV - 7V V - 0V, V + = 5V, VCM = 5V Small-Signal Voltage Gain (Note 10) 1.5 IOUT OVERDRIVE = 50mV IOUT = 10mA, V + = 5V, VOVERDRIVE = 50mV V + - 0.5 V + - 0.7 Output Voltage Swing LOW IOUT = - 1mA, VOVERDRIVE = 50mV IOUT = - 10mA, VOVERDRIVE = 50mV Positive Supply Current (Per Comparator) V + = 5V, VOVERDRIVE = 1V VOH Output Voltage Swing HIGH VOL I+ = 1mA, V + = 5V, V V + - 0.2 Negative Supply Current (Per Comparator) V + = 5V, VOVERDRIVE = 1V 0.20 0.35 0.4 0.5 V V 5 6.5 8.0 mA mA 3 4.0 4.5 mA mA VIH Latch Pin High Input Voltage VIL Latch Pin Low Input Voltage 2.4 V = V+ IIL Latch Pin Current VLATCH tPD Propagation Delay (Note 6) VIN = 100mV, VOVERDRIVE = 20mV VIN = 100mV, VOVERDRIVE = 20mV VIN = 100mV, VOVERDRIVE = 5mV UNITS V V V + - 0.4 I- MAX 8.0 0.8 V 10 A 11.0 12.5 ns ns ns 9.0 tPD Differential Propagation Delay (Note 6) VIN = 100mV, VOVERDRIVE = 20mV tr Output Rise Time 10% to 90% 4 ns tf Output Fall Time 90% to 10% 4 ns tLPD Latch Propagation Delay (Note 7) 8 ns tSU Latch Setup Time (Note 7) 1.5 ns tH Latch Hold Time (Note 7) 0 ns tDPW Minimum Latch Disable Pulse Width (Note 7) 8 ns fMAX Maximum Toggle Frequency VIN = 100mVP-P Sine Wave 65 MHz tJITTER Output Timing Jitter VIN = 630mVP-P (0dBm) Sine Wave, f = 30MHz 15 psRMS 0.5 3 ns The denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. V + = 5V, V - = - 5V, VCM = 0V, VLATCH = 0.8V, CLOAD = 10pF, VOVERDRIVE = 20mV, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS V+ Positive Supply Voltage Range 2.4 7 V V- Negative Supply Voltage Range (Note 3) -7 0 V VOS Input Offset Voltage (Note 4) 3 4 mV mV mV mV VOS/T Input Offset Voltage Drift IOS Input Offset Current RS = 50, VCM = 0V RS = 50, VCM = 0V RS = 50, VCM = - 5V RS = 50, VCM = 5V 0.5 0.7 1 0.1 1 2 A A - 1.5 2 5 A A IB Input Bias Current (Note 5) V/C 5 -7 - 15 3 LT1713/LT1714 ELECTRICAL CHARACTERISTICS The denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. V + = 5V, V - = - 5V, VCM = 0V, VLATCH = 0.8V, CLOAD = 10pF, VOVERDRIVE = 20mV, unless otherwise specified. SYMBOL PARAMETER CONDITIONS VCM Input Voltage Range CMRR Common Mode Rejection Ratio - 5V VCM 5V Positive Power Supply Rejection Ratio 2.4V V + PSRR+ PSRR- Negative Power Supply Rejection Ratio MIN TYP - 5.1 70 62 60 dB dB 68 65 80 dB dB 65 60 80 dB dB 1.5 3 V/mV 4.5 4.3 4.8 4.6 7V, VCM = - 5V - 7V V - 0V, VCM = 5V Small-Signal Voltage Gain (Note 10) 1V VOUT 4V, RL = VOH Output Voltage Swing HIGH (Note 8) IOUT = 1mA, VOVERDRIVE = 50mV IOUT = 10mA, VOVERDRIVE = 50mV VOL Output Voltage Swing LOW (Note 8) IOUT = - 1mA, VOVERDRIVE = 50mV IOUT = - 10mA, VOVERDRIVE = 50mV I+ Positive Supply Current (Per Comparator) VOVERDRIVE = 1V 5.1 VOVERDRIVE = 1V V V V 0.20 0.35 0.4 0.5 V V 5.5 7.5 9.0 mA mA 3.5 4.5 5.0 mA mA Negative Supply Current (Per Comparator) UNITS AV I- MAX VIH Latch Pin High Input Voltage VIL Latch Pin Low Input Voltage 0.8 V IIL Latch Pin Current VLATCH = V + 10 A tPD Propagation Delay (Note 6) VIN = 100mV, VOVERDRIVE = 20mV VIN = 100mV, VOVERDRIVE = 20mV VIN = 100mV, VOVERDRIVE = 5mV 10 12 ns ns ns 2.4 V 7 8.5 tPD Differential Propagation Delay (Note 6) VIN = 100mV, VOVERDRIVE = 20mV tr Output Rise Time 10% to 90% 4 ns tf Output Fall Time 90% to 10% 4 ns tLPD Latch Propagation Delay (Note 7) 8 ns tSU Latch Setup Time (Note 7) 1.5 ns tH Latch Hold Time (Note 7) 0 ns tDPW Minimum Latch Disable Pulse Width (Note 7) 8 ns fMAX Maximum Toggle Frequency VIN = 100mVP-P Sine Wave 65 MHz tJITTER Output Timing Jitter VIN = 630mVP-P (0dBm) Sine Wave, f = 30MHz 15 psRMS Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT1713C/LT1714C are guaranteed to meet specified performance from 0C to 70C. They are designed, characterized and expected to meet specified performance from - 40C to 85C but are not tested or QA sampled at these temperatures. The LT1713I/LT1714I are guaranteed to meet specified performance from - 40C to 85C. Note 3: The negative supply should not be greater than the ground pin voltages and the maximum voltage across the positive and negative supplies should not be greater than 12V. Note 4: Input offset voltage (VOS) is defined as the average of the two voltages measured by forcing first one output, then the other to V +/2. Note 5: Input bias current (IB) is defined as the average of the two input currents. 4 0.5 3 ns Note 6: Propagation delay (tPD) is measured with the overdrive added to the actual VOS. Differential propagation delay is defined as: tPD = tPD+ - tPD-. Load capacitance is 10pF. Due to test system requirements, the LT1713/LT1714 propagation delay is specified with a 1k load to ground for 5V supplies, or to mid-supply for 2.7V or 5V single supplies. Note 7: Latch propagation delay (tLPD) is the delay time for the output to respond when the latch pin is deasserted. Latch setup time (tSU) is the interval in which the input signal must remain stable prior to asserting the latch signal. Latch hold time (tH) is the interval after the latch is asserted in which the input signal must remain stable. Latch disable pulse width (tDPW) is the width of the negative pulse on the latch enable pin that latches in new data on the data inputs. LT1713/LT1714 ELECTRICAL CHARACTERISTICS Note 8: Output voltage swings are characterized and tested at V + = 5V and V - = 0V. They are designed and expected to meet these same specifications at V - = - 5V. Note 9: The input voltage range is tested under the more demanding conditions of V + = 5V and V - = -5V. The LT1713/LT1714 are designed and expected to meet these specifications at V - = 0V. Note 10: The LT1713/LT1714 voltage gain is tested at V + = 5V and V - = - 5V only. Voltage gain at single supply V + = 5V and V + = 2.7V is guaranteed by design and correlation. Note 11: Input offset voltage over temperature at V + = 2.7V is guaranteed by design and characterization. U W TYPICAL PERFOR A CE CHARACTERISTICS Input Offset Voltage vs Temperature 1.0 0.5 0 - 0.5 - 1.0 - 1.5 14 16 12 14 tPD+ 10 tPD- 8 6 TA = 25C V + = 5V V - = 0V VCM = 2.5V VOD = 20mV VSTEP = 100mV 4 2 - 2.0 - 2.5 -50 -25 50 0 75 25 TEMPERATURE (C) 100 0 125 PROPAGATION DELAY (ns) 1.5 V+ = 5V V - = 0V VCM = 2.5V Propagation Delay vs Temperature 0 20 80 60 100 40 LOAD CAPACITANCE (pF) 171314 G01 10.0 9.5 9.5 tPD 8.0 tPD- 7.5 TA = 25C V + = 5V V - = 0V VOD = 20mV VSTEP = 100mV CLOAD = 10pF 7.0 6.5 6.0 - 0.5 0.5 1.5 3.5 4.5 2.5 INPUT COMMON MODE (V) 5.5 171314 G04 PROPAGATION DELAY (ns) PROPAGATION DELAY (ns) 10.0 + 10 tPD+ 8 tPD- 6 4 0 -50 -25 120 0 25 50 75 100 TA = 25C V - = 0V VCM = 2.5V VOD = 20mV VSTEP = 100mV CLOAD = 10pF 9.0 8.5 8.0 tPD+ tPD- 7.5 7.0 6.5 6.0 0 2 10 12 8 6 POSITIVE SUPPLY VOLTAGE (V) 4 125 TEMPERATURE (C) Propagation Delay vs Positive Supply Voltage 8.5 12 171314 G03 171314 G02 Propagation Delay vs Input Common Mode Voltage 9.0 V+ = 5V V - = 0V VCM = 2.5V VOD = 20mV VSTEP = 100mV CLOAD = 10pF 2 14 171314 G05 POSITIVE SUPPLY CURRENT (PER COMPARATOR) (mA) INPUT OFFSET VOLTAGE (mV) 2.0 PROPAGATION DELAY (ns) 2.5 Propagation Delay vs Load Capacitance Positive Supply Current vs Positive Supply Voltage 7.0 TA = 25C VIN = 100mV 6.5 IOUT = 0 V - = -5V 6.0 V - = 0V 5.5 5.0 4.5 4.0 2 6 8 10 4 POSITIVE SUPPLY VOLTAGE (V) 12 171314 G06 5 LT1713/LT1714 U W 40 TA = 25C + 35 V - = 5V V = 0V 30 CLOAD = 10pF 25 20 15 10 5 0 20 10 30 SWITCHING FREQUENCY (MHz) 40 4.0 3.8 3.6 3.2 V + = 2.7V 3.0 2.8 2.6 2.4 OUTPUT VOLTAGE (V) INPUT BIAS CURRENT (A) 0 -1 -2 -3 -4 2.2 -5 2.0 -6 0 -3 -4 -2 -5 -6 -1 NEGATIVE SUPPLY VOLTAGE (V) -7 -1 -2 -3 50 75 100 125 TEMPERATURE (C) 1 2 0 3 4 5 INPUT COMMON MODE VOLTAGE (V) Output Low Voltage vs Sink Current 5.0 1.0 4.9 0.9 4.8 0.8 4.7 4.6 4.5 4.4 4.3 T = 25C 4.2 VA+ = 5V - 4.1 V = 0V VIN = 100mV 4.0 1 0.01 0.1 LOADING SOURCE CURRENT (mA) 171314 G10 TA = 25C V + = 5V V - = 0V VIN = 100mV 0.7 0.6 0.5 0.4 0.3 0.2 0.1 10 0 0.01 171314 G11 Output Timing Jitter vs Switching Frequency 6 171314 G09 Output High Voltage vs Source Current -1 25 1 TA = 25C V + = 5V V - = 0V VIN = 0mV 171314 G08 V+ = 5V V - = 0V VCM = 2.5V 0 2 V + = 5V 3.4 Input Bias Current vs Temperature -4 -50 -25 3 TA = 25C VIN = 100mV IOUT = 0 171314 G07 0 Input Bias Current vs Input Common Mode Voltage OUTPUT VOLTAGE (V) 0 Negative Supply Current vs Negative Supply Voltage INPUT BIAS CURRENT (A) Positive Supply Current vs Switching Frequency NEGATIVE SUPPLY CURRENT (PER COMPARATOR) (mA) POSITIVE SUPPLY CURRENT (PER COMPARATOR) (mA) TYPICAL PERFOR A CE CHARACTERISTICS 1 0.1 LOADING SINK CURRENT (mA) 10 171314 G12 Output Rising Edge, 5V Supply Output Falling Edge, 5V Supply 200 TA = 25C V + = 5V V - = 0V VCM = 2.5V VIN = 630mVP-P (0dBm) SINE WAVE OUTPUT TIMING JITTER (psRMS) 180 160 140 120 VIN VIN VOUT VOUT 100 80 60 40 171314 G14 20 0 0 20 40 60 FREQUENCY (MHz) 80 171314 G13 6 171314 G15 LT1713/LT1714 U U U PI FU CTIO S LT1713 V + (Pin 1): Positive Supply Voltage, Usually 5V. GND (Pin 6): Ground Supply Voltage, Usually 0V. + IN (Pin 2): Noninverting Input. Q (Pin 7): Noninverting Output. - IN (Pin 3): Inverting Input. Q (Pin 8): Inverting Output. V - (Pin 4): Negative Supply Voltage, Usually 0V or - 5V. LATCH ENABLE (Pin 5): Latch Enable Input. With a logic high the output is latched. LT1714 - IN A (Pin 1): Inverting Input of A Channel Comparator. + IN A (Pin 2): Noninverting Input of A Channel Comparator. V - (Pins 3, 6): Negative Supply Voltage, Usually - 5V. Pins 3 and 6 should be connected together externally. V + (Pins 4, 5): Positive Supply Voltage, Usually 5V. Pins 4 and 5 should be connected together externally. + IN B (Pin 7): Noninverting Input of B Channel Comparator. Q B (Pin 11): Noninverting Output of B Channel Comparator. Q B (Pin 12): Inverting Output of B Channel Comparator. Q A (Pin 13): Inverting Output of A Channel Comparator. Q A (Pin 14): Noninverting Output of A Channel Comparator. - IN B (Pin 8): Inverting Input of B Channel Comparator. GND (Pin 15): Ground Supply Voltage of A Channel Comparator, Usually 0V LATCH ENABLE B (Pin 9): Latch Enable Input of B Channel Comparator. With a logic high, the B output is latched. LATCH ENABLE A (Pin 16): Latch Enable Input of A Channel Comparator. With a logic high, the A output is latched. GND (Pin 10): Ground Supply Voltage of B Channel Comparator, Usually 0V. 7 LT1713/LT1714 U W U U APPLICATIO S I FOR ATIO Common Mode Considerations Latch Pin Dynamics The LT1713/LT1714 are specified for a common mode range of - 5.1V to 5.1V on a 5V supply, or a common mode range of - 0.1V to 5.1V on a single 5V supply. A more general consideration is that the common mode range is from 100mV below the negative supply to 100mV above the positive supply, independent of the actual supply voltage. The criteria for common mode limit is that the output still responds correctly to a small differential input signal. The internal latches of the LT1713/LT1714 comparators retain the input data (output latched) when their respective latch pin goes high. The latch pin will float to a low state when disconnected, but it is better to ground the latch when a flow-through condition is desired. The latch pin is designed to be driven with either a TTL or CMOS output. It has built-in hysteresis of approximately 100mV, so that slow moving or noisy input signals do not impact latch performance. For the LT1714, if only one of the comparators is being used at a given time, it is best to latch the second comparator to avoid any possibility of interactions between the two comparators in the same package. When either input signal falls outside the common mode limit, the internal PN diode formed with the substrate can turn on resulting in significant current flow through the die. Schottky clamp diodes between the inputs and the supply rails speed up recovery from excessive overdrive conditions by preventing these substrate diodes from turning on. Input Bias Current Input bias current is measured with the outputs held at 2.5V with a 5V supply voltage. As with any rail-to-rail differential input stage, the LT1713/LT1714 bias current flows into or out of the device depending upon the common mode level. The input circuit consists of an NPN pair and a PNP pair. For inputs near the negative rail, the NPN pair is inactive, and the input bias current flows out of the device; for inputs near the positive rail, the PNP pair is inactive, and these currents flow into the device. For inputs far enough away from the supply rails, the input bias current will be some combination of the NPN and PNP bias currents. As the differential input voltage increases, the input current of each pair will increase for one of the inputs and decrease for the other input. Large differential input voltages result in different input currents as the input stage enters various regions of operation. To reduce the influence of these changing input currents on system operation, use a low source resistance. 8 High Speed Design Techniques A substantial amount of design effort has made the LT1713/LT1714 relatively easy to use. As with most high speed comparators, careful attention to PC board layout and design is important in order to prevent oscillations. The most common problem involves power supply bypassing which is necessary to maintain low supply impedance. Resistance and inductance in supply wires and PC traces can quickly build up to unacceptable levels, thereby allowing the supply voltages to move as the supply current changes. This movement of the supply voltages will often result in improper operation. In addition, adjacent devices connected through an unbypassed supply can interact with each other through the finite supply impedances. Bypass capacitors furnish a simple solution to this problem by providing a local reservoir of energy at the device, thus keeping supply impedance low. Bypass capacitors should be as close as possible to the LT1713/LT1714 supply pins. A good high frequency capacitor, such as a 0.1F ceramic, is recommended in parallel with a larger capacitor, such as a 4.7F tantalum. LT1713/LT1714 U W U U APPLICATIO S I FOR ATIO Poor trace routes and high source impedances are also common sources of problems. Keep trace lengths as short as possible and avoid running any output trace adjacent to an input trace to prevent unnecessary coupling. If output traces are longer than a few inches, provide proper termination impedances (typically 100 to 400) to eliminate any reflections that may occur. Also keep source impedances as low as possible, preferably much less than 1k. The input and output traces should also be isolated from one another. Power supply traces can be used to achieve this isolation as shown in Figure 1, a typical topside layout of the LT1713/LT1714 on a multilayer PC board. Shown is the topside metal etch including traces, pin escape vias and the land pads for a GN16 LT1713/LT1714 and its adjacent X7R 0805 bypass capacitors. The V +, V - and GND traces all shield the inputs from the outputs. Although the two V - pins are connected internally, they should be shorted together externally as well in order for both to function as shields. The same is true for the two V + pins. The two GND pins are not connected internally, but in most applications they are both connected directly to the ground plane. 1714 F01 Figure 1. Typical LT1714 Topside Metal for Multilayer PCB Layout 9 LT1713/LT1714 U W U U APPLICATIO S I FOR ATIO Hysteresis Another useful technique to avoid oscillations is to provide positive feedback, also known as hysteresis, from the output to the input. Increased levels of hysteresis, however, reduce the sensitivity of the device to input voltage levels, so the amount of positive feedback should be tailored to particular system requirements. The + 50 LT1713 VIN Q 50 VIN+ + - 50k 100k 50k Q VIN LT1713/LT1714 are completely flexible regarding the application of hysteresis, due to rail-to-rail inputs and the complementary outputs. Specifically, feedback resistors can be connected from one or both outputs to their corresponding inputs without regard to common mode considerations. Figure 2 shows several configurations. LT1713 VREF Q V + = 5V V - = -5V VHYST = 5mV (ALL 3 CASES) VIN- - 50 50 LT1713 - Q Q Figure 2. Various Configurations for Introducing Hysteresis 10 Q + 100k 171314 F02 LT1713/LT1714 U TYPICAL APPLICATIO S Simultaneous Full Duplex 75Mbaud Interface with Only Two Wires The circuit of Figure 3 shows a simple, fully bidirectional, differential 2-wire interface that gives good results to 75Mbaud, using the LT1714. Eye diagrams under conditions of unidirectional and bidirectional communication are shown in Figures 4 and 5. Although not as pristine as the unidirectional performance of Figure 4, the performance under simultaneous bidirectional operation is still excellent. Because the LT1714 input voltage range extends 100mV beyond both supply rails, the circuit works with a full 3V (one whole VS up or down) of ground potential difference. The circuit works well with the resistor values shown, but other sets of values can be used. The starting point is the characteristic impedance, ZO, of the twisted-pair cable. The input impedance of the resistive network should match the characteristic impedance and is given by: RIN = 2 * RO * R1||(R2 + R3) RO + 2 * [R1||(R2 + R3)] This comes out to 120 for the values shown. The Thevenin equivalent source voltage is given by: (R2 + R3 - R1) (R2 + R3 + R1) RO * RO + 2 * [R1||(R2 + R3)] VTH = VS * 750k 750k 3V 3V 4 + 14 1/2 LE LT1714 RxD 13 16 3 - 2 2 1 1 3V TxD 49.9 8 - 3 3V 100k 7 + 14 1/2 LT1714 LE 15 750k 49.9 4 + R2A 2.55k 5 750k 3V 100k R1C 499 R1A 499 R2C 2.55k 100k 3V 7 49.9 1/2 LT1714 8 LE - 12 6 10 49.9 5 + 11 11 1/2 LT1714 - LE 10 12 6 9 R3A 124 ROA 140 R3B 124 R1B 499 R2B 2.55k 6-FEET TWISTED PAIR ZO 120 DIODES: BAV99 x4 15 RxD 16 13 ROB 140 R3C 124 R1D 499 R3D 124 R2D 2.55k TxD 9 100k 171314 F03 Figure 3. 75Mbaud Full Duplex Interface on Two Wires 11 LT1713/LT1714 U TYPICAL APPLICATIO S 171112 F05 171112 F04 Figure 5. Performance When Operated Simultaneous Bidirectionally (Full Duplex). Crosstalk Appears as Noise. Eye is Slightly Shut But Performance is Still Excellent Figure 4. Performance of Figure 3's Circuit When Operated Unidirectionally. Eye is Wide Open This amounts to an attenuation factor of 0.0978 with the values shown. (The actual voltage on the lines will be cut in half again due to the 120 ZO.) The reason this attenuation factor is important is that it is the key to deciding the ratio between the R2-R3 resistor divider in the receiver path. This divider allows the receiver to reject the large signal of the local transmitter and instead sense the attenuated signal of the remote transmitter. Note that in the above equations, R2 and R3 are not yet fully determined because they only appear as a sum. This allows the designer to now place an additional constraint on their values. The R2-R3 divide ratio should be set to equal half the attenuation factor mentioned above or: The LT1713 is set up as a crystal oscillator. The varactor diode is biased from the tuning input. The tuning network is arranged so a 0V to 5V drive provides a reasonably symmetric, broad tuning range around the 14.31818MHz center frequency. The indicated selected capacitor sets tuning bandwidth. It should be picked to complement loop response in phase locking applications. Figure 6 is a plot of tuning input voltage versus frequency deviation. Tuning deviation from the 4 x NTSC 14.31818MHz center frequency exceeds 240ppm for a 0V to 5V input. 1 Using the design value of R2 + R3 = 2.653k rather than the implementation value of 2.55k + 124 = 2.674k. R3/R2 = 1/2 * 0.09761. Voltage-Tunable Crystal Oscillator The front page application is a variant of a basic crystal oscillator that permits voltage tuning of the output frequency. Such voltage-controlled crystal oscillators (VCXO) are often employed where slight variation of a stable carrier is required. This example is specifically intended to provide a 4 x NTSC sub-carrier tunable oscillator suitable for phase locking. 12 14.3217MHz 8 FREQUENCY DEVIATION (kHz) Having already designed R2 + R3 to be 2.653k (by allocating input impedance across RO, R1 and R2 + R3 to get the requisite 120), R2 and R3 then become 2529 and 123.5 respectively. The nearest 1% value for R2 is 2.55k and that for R3 is 124. 9 7 6 5 14.31818MHz 4 3 2 1 14.314.0MHz 0 0 1 3 2 INPUT VOLTAGE (V) 5 4 171112 F06 Figure 6. Control Voltage vs Output Frequency for the First Page Application Circuit. Tuning Deviation from Center Frequency Exceeds 240ppm LT1713/LT1714 U TYPICAL APPLICATIO S 1MHz Series Resonant Crystal Oscillator with Square and Sinusoid Outputs Figure 7 shows a classic 1MHz series resonant crystal oscillator. At series resonance, the crystal is a low impedance and the positive feedback connection is what brings about oscillation at the series resonant frequency. The RC feedback around the other path ensures that the circuit does not find a stable DC operating point and refuse to oscillate. The comparator output is a 1MHz square wave (top trace of Figure 8) with jitter measured at better than 28psRMS on a 5V supply and 40psRMS on a 3V supply. At Pin 2 of the comparator, on the other side of the crystal, is a clean sine wave except for the presence of the small high frequency glitch (middle trace of Figure 8). This glitch is R10 1k 1MHz AT-CUT R4 210 VS R1 1k R2 1k caused by the fast edge of the comparator output feeding back through crystal capacitance. Amplitude stability of the sine wave is maintained by the fact that the sine wave is basically a filtered version of the square wave. Hence, the usual amplitude control loops associated with sinusoidal oscillators are not necessary.2 The sine wave is filtered and buffered by the fast, low noise LT1806 op amp. To remove the glitch, the LT1806 is configured as a bandpass filter with a Q of 5 and unity-gain center frequency of 1MHz, with its output shown as the bottom trace of Figure 8. Distortion was measured at - 70dBc and - 60dBc on the second and third harmonics, respectively. 2 Amplitude will be a linear function of comparator output swing, which is supply dependent and therefore adjustable. The important difference here is that any added amplitude stabilization or control loop will not be faced with the classical task of avoiding regions of nonoscillation versus clipping. C4 100pF R5 6.49k C5 100pF R6 162 C3 100pF 2 VS 2 3 1 + - 7 LT1713 LE 5 VS 4 6 SQUARE R9 2k 3 C2 0.1F R8 2k VS 8 - R7 15.8k 7 6 LT1806 + SINE 1 4 171314 F07 R3 1k C1 0.1F Figure 7. LT1713 Comparator is Configured as a Series Resonant Xtal Oscillator. LT1806 Op Amp is Configured in a Q = 5 Bandpass with fC = 1MHz 3V/DIV 1V/DIV 1V/DIV 200ns/DIV 171112 F08 Figure 8. Oscillator Waveforms with VS = 3V. Top is Comparator Output. Middle is Xtal Feedback to Pin 2 at LT1713 (Note the Glitches). Bottom is Buffered, Inverted and Bandpass Filtered with a Q = 5 by LT1806 13 LT1713/LT1714 U PACKAGE DESCRIPTIO Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 0.004* (3.00 0.102) 8 7 6 5 0.118 0.004** (3.00 0.102) 0.193 0.006 (4.90 0.15) 1 2 3 4 0.043 (1.10) MAX 0.007 (0.18) 0 - 6 TYP 0.021 0.006 (0.53 0.015) SEATING PLANE 0.009 - 0.015 (0.22 - 0.38) 0.0256 (0.65) BSC * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE 14 0.034 (0.86) REF 0.005 0.002 (0.13 0.05) MSOP (MS8) 1100 LT1713/LT1714 U PACKAGE DESCRIPTIO Dimensions in inches (millimeters) unless otherwise noted. GN Package 16-Lead Plastic SSOP (Narrow 0.150) (LTC DWG # 05-08-1641) 0.189 - 0.196* (4.801 - 4.978) 16 15 14 13 12 11 10 9 0.229 - 0.244 (5.817 - 6.198) 0.150 - 0.157** (3.810 - 3.988) 1 0.015 0.004 x 45 (0.38 0.10) 0.007 - 0.0098 (0.178 - 0.249) 0.009 (0.229) REF 2 3 4 5 6 7 0.053 - 0.068 (1.351 - 1.727) 8 0.004 - 0.0098 (0.102 - 0.249) 0 - 8 TYP 0.016 - 0.050 (0.406 - 1.270) 0.008 - 0.012 (0.203 - 0.305) 0.0250 (0.635) BSC * DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE GN16 (SSOP) 1098 U TYPICAL APPLICATIO Rail-to-Rail Pulse Width Modulator Using the LT1714 Binary modulation schemes are used in order to improve efficiency and reduce physical circuit size. They do this by reducing the power dissipation in the output driver transistors. In a normal Class A or Class AB amplifier, voltage drop and current flow exist simultaneously in the output transistors and power losses proportional to V * I occur. In a binary modulation scheme, the output transistors, whether bipolar or FET, are switched hard-on and hard-off so that voltage drops do not occur simultaneously with current flow. The circuit of Figure 9 shows an example of a binary modulation scheme, in this case pulse width modulation. The LT1809 is configured as an integrator in order to generate nice linear rail-to-rail voltage ramps. The polarity of the ramp is determined by the output of the LT1714's comparator A into R4. The heavy hysteresis of R1 around the LT1714's comparator A combined with the feedback of the LT1809 force the devices to perpetually reverse each other, resulting in a 1MHz triangle wave. This constitutes the usual first half of any pulse width modulator, but the Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1713/LT1714 U TYPICAL APPLICATIO forte of this particular implementation is that it is rail-torail allowing a full-scale analog input. Once the triangle wave is achieved, the remainder of the pulse width modulator is easy, and is constituted by doing a simple comparison using the second half of the LT1714. The triangle wave and the relatively slow moving analog signal (the one to be modulated or to do the modulation, depending on how you look at it) are fed into the inputs of comparator B, whose output is then the PWM representation of the analog input voltage. The higher the analog input voltage, the wider the output pulse. The time averaged output level is thus proportional to the analog input voltage. This binary output can then be fed into power transistors with direct control over motor or speaker winding current, for R1 26.1 V+ R2 2k R3 2k V+ 2 4 + A 1/2 LT1714 1 - 14 R4 13 499 2 16 - 3 V+ 7 LT1809 3 R5 1k + C2 0.01F C4 0.001F 6 4 R6 1k V + = 2.7V TO 7V 21 C3 100pF 1MHz TRIANGLE WAVE 16kHz ANALOG FILTER FOR LINEARITY MEASUREMENT V+ 7 V+ 1 15 The linearity of the pulse width modulated signal can easily be ascertained by putting a simple 2-pole RC filter at the output (as shown in Figure 9). This demodulates the signal which can then be viewed and compared with the original input signal on an oscilloscope. Using a spectrum analyzer and a 1kHz reference signal, this circuit's distortion products were measured as better than - 50dBc (0.3%) to about 3.5VP-P, degrading to - 30dBc (3%) as the circuit clips at 5VP-P on a single 5V supply. ANTIALIASING 1k FILTER ANALOG INPUT C1 500pF example, with their inherent lowpass characteristic. Care must be taken to avoid cross conduction in the output power transistors. 5 + B 1/2 LT1714 8 - 1k 11 12 10 10k C5 0.01F C6 0.001F 9 6 171314 F09 COMPLEMENTARY 1MHz PWM OUTPUTS Figure 9. Rail-to-Rail 1MHz Pulse Width Modulator RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1016 UltraFast Precision Comparator Industry Standard 10ns Comparator LT1116 12ns Single Supply Ground Sensing Comparator Single Supply Version of the LT1016 LT1394 7ns, UltraFast Single Supply Comparator 6mA Single Supply Comparator LT1671 60ns, Low Power, Single Supply Comparator 450A Single Supply Comparator LT1711/LT1712 Single/Dual, 4.5ns, 3V/5V/5V Rail-to-Rail Comparators Faster Versions of LT1713/LT1714 LT1719 4.5ns, Single Supply 3V/5V Comparator 4mA Comparator with Rail-to-Rail Outputs LT1720/LT1721 Dual/Quad, 4.5ns, Single Supply Comparator Dual/Quad Version of the LT1719 16 Linear Technology Corporation 171314f LT/TP 0501 4K * PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 FAX: (408) 434-0507 www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 2000