1
LT1372/LT1377
500kHz and 1MHz
High Efficiency
1.5A Switching Regulators
Boost Regulators
CCFL Backlight Driver
Laptop Computer Supplies
Multiple Output Flyback Supplies
Inverting Supplies
The LT
®
1372/LT1377 are monolithic high frequency
switching regulators. They can be operated in all standard
switching configurations including boost, buck, flyback,
forward, inverting and “Cuk.” A 1.5A high efficiency switch
is included on the die, along with all oscillator, control and
protection circuitry. All functions of the LT1372/LT1377
are integrated into 8-pin SO/PDIP packages.
The LT1372/LT1377 typically consumes only 4mA quies-
cent current and has higher efficiency than previous parts.
High frequency switching allows for very small inductors
to be used. All surface mount components consume less
than 0.5 square inch of board space.
New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an exter-
nal logic level source. A logic low on the shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation tech-
niques. Nonlinear error amplifier transconductance re-
duces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external compo-
nents during overload conditions.
Faster Switching with Increased Efficiency
Uses Small Inductors: 4.7µH
All Surface Mount Components
Only 0.5 Square Inch of Board Space
Low Minimum Supply Voltage: 2.7V
Quiescent Current: 4mA Typ
Current Limited Power Switch: 1.5A
Regulates Positive or Negative Outputs
Shutdown Supply Current: 12µA Typ
Easy External Synchronization
8-Pin SO or PDIP Packages
, LTC and LT are registered trademarks of Linear Technology Corporation.
12V Output Efficiency
OUTPUT CURRENT (A)
0.01
50
EFFICIENCY (%)
60
70
80
90
0.1 1
LT1372 • TA02
100 V
IN
= 5V
5V-to-12V Boost Converter
LT1372/LT1377
V
IN
V
C
5V
1
2
8
5
4
6, 7
GND
FB
LT1372 • TA01
V
SW
S/S
L1*
4.7µH
C1**
22µFC4**
22µF
C2
0.047µFC3
0.0047µF
R3
2k
R2
6.19k
1%
R1
53.6k
1%
V
OUT
12V
D1
MBRS120T3
ON
OFF *FOR LT1372 USE 10µH
COILCRAFT DO1608-472 (4.7µH) OR
COILCRAFT DT3316-103 (10µH) OR
SUMIDA CD43-4R7 (4.7µH) OR
SUMIDA CD73-100KC (10µH) OR
**AVX TPSD226M025R0200
L1
4.7µH
10µH
I
OUT
(LT1377)
0.25A
0.35A
I
OUT
(LT1372)
NA
0.29A
MAX I
OUT
+ +
FEATURES
DESCRIPTIO
U
APPLICATIO S
U
TYPICAL APPLICATIO
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2
LT1372/LT1377
(Note 1)
Supply Voltage ....................................................... 30V
Switch Voltage
LT1372/LT1377 .................................................. 35V
LT1372HV .......................................................... 42V
S/S Pin Voltage....................................................... 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms)............................................. ±10V
Operating Junction Temperature Range
Commercial ........................................ 0°C to 125°C*
Industrial ......................................... 40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Storage Temperature Range ................ 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART NUMBER
LT1372CN8
LT1372HVCN8
LT1372CS8
LT1372HVCS8
LT1372IN8
LT1372HVIN8
LT1372IS8
LT1372HVIS8
LT1377CS8
LT1377IS8
1
2
3
4
8
7
6
5
TOP VIEW
V
C
FB
NFB
S/S
V
SW
GND
GND S
V
IN
N8 PACKAGE
8-LEAD PDIP S8 PACKAGE
8-LEAD PLASTIC SO
T
JMAX
= 125°C, θ
JA
= 100°C/ W (N8)
T
JMAX
= 125°C, θ
JA
= 120°C/ W (S8)
S8 PART MARKING
1372
1372I 1377
1377I
1372H
1372HI
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
V
REF
Reference Voltage Measured at Feedback Pin 1.230 1.245 1.260 V
V
C
= 0.8V 1.225 1.245 1.265 V
I
FB
Feedback Input Current V
FB
= V
REF
250 550 nA
900 nA
Reference Voltage Line Regulation 2.7V V
IN
25V, V
C
= 0.8V 0.01 0.03 %/V
V
NFB
Negative Feedback Reference Voltage Measured at Negative Feedback Pin 2.540 2.490 2.440 V
Feedback Pin Open, V
C
= 0.8V 2.570 2.490 2.410 V
I
NFB
Negative Feedback Input Current V
NFB
= V
NFR
–45 –30 15 µA
Negative Feedback Reference Voltage 2.7V V
IN
25V, V
C
= 0.8V 0.01 0.05 %/V
Line Regulation
g
m
Error Amplifier Transconductance I
C
= ±25µA 1100 1500 1900 µmho
700 2300 µmho
Error Amplifier Source Current V
FB
= V
REF
– 150mV, V
C
= 1.5V 120 200 350 µA
Error Amplifier Sink Current V
FB
= V
REF
+ 150mV, V
C
= 1.5V 1400 2400 µA
Error Amplifier Clamp Voltage High Clamp, V
FB
= 1V 1.70 1.95 2.30 V
Low Clamp, V
FB
= 1.5V 0.25 0.40 0.52 V
A
V
Error Amplifier Voltage Gain 500 V/V
V
C
Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V
f Switching Frequency 2.7V V
IN
25V
LT1372 450 500 550 kHz
0°C T
J
125°C430 500 580 kHz
40°C T
J
< 0°C (I Grade) 400 580 kHz
LT1377 0.90 1 1.10 MHz
0°C T
J
125°C0.86 1 1.16 MHz
40°C T
J
< 0°C (I Grade) 0.80 1.16 MHz
The denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
Consult factory for parts specified with wider operating temperature ranges.
*Units shipped prior to Date Code 9552 are rated at 100°C maximum
operating temperature.
ABSOLUTE AXI U RATI GS
WWWU
PACKAGE/ORDER I FOR ATIO
UU
W
ELECTRICAL CHARACTERISTICS
3
LT1372/LT1377
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Maximum Switch Duty Cycle 85 95 %
Switch Current Limit Blanking Time 130 260 ns
BV Output Switch Breakdown Voltage LT1372/LT1377 35 47 V
LT1372HV
0°C T
J
125°C42 47 V
40°C T
J
< 0°C (I Grade) 40 V
V
SAT
Output Switch “On” Resistance I
SW
= 1A 0.5 0.8
I
LIM
Switch Current Limit Duty Cycle = 50% 1.5 1.9 2.7 A
Duty Cycle = 80% (Note 2) 1.3 1.7 2.5 A
I
IN
Supply Current Increase During Switch On-Time 15 25 mA/A
I
SW
Control Voltage to Switch Current 2A/V
Transconductance
Minimum Input Voltage 2.4 2.7 V
I
Q
Supply Current 2.7V V
IN
25V 4 5.5 mA
Shutdown Supply Current 2.7V V
IN
25V, V
S/S
0.6V
0°C T
J
125°C12 30 µA
40°C T
J
< 0°C (I Grade) 50 µA
Shutdown Threshold 2.7V V
IN
25V 0.6 1.3 2 V
Shutdown Delay 51225µs
S/S Pin Input Current 0V V
S/S
5V –10 15 µA
Synchronization Frequency Range LT1372 600 800 kHz
LT1377 1.2 1.6 MHz
Switch Saturation Voltage
vs Switch Current
TEMPERATURE (°C)
–50
1.8
INPUT VOLTAGE (V)
2.0
2.2
2.4
2.6
050
100 150
LT1372 • G03
2.8
3.0
–25 25 75 125
Minimum Input Voltage
vs Temperature
DUTY CYCLE (%)
0
SWITCH CURRENT LIMIT (A)
1.0
2.0
3.0
0.5
1.5
2.5
20 40 60 80
LT1372 • G02
10010
030 50 70 90
25°C AND
125°C
–55°C
Switch Current Limit
vs Duty Cycle
TYPICAL PERFORMANCE CHARACTERISTICS
UW
SWITCH CURRENT (A)
0
SWITCH SATURATION VOLTAGE (V)
0.6
0.8
1.0
1.6
LT1372 • G01
0.4
0.2
0.5
0.7
0.9
0.3
0.1
00.4 0.8 1.2 2.0
1.4
0.2 0.6 1.0 1.8
100°C
150°C
25°C
–55°C
Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.
The denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
ELECTRICAL CHARACTERISTICS
Note 2: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by I
LIM
= 0.667 (2.75 – DC).
4
LT1372/LT1377
Error Amplifier Output Current
vs Feedback Pin Voltage
Shutdown Delay and Threshold
vs Temperature
TEMPERATURE (°C)
–50
0
SHUTDOWN DELAY (µs)
SHUTDOWN THRESHOLD (V)
2
6
8
10
20
14
050 75
LT1372 • G04
4
16
18
12
0
0.2
0.6
0.8
1.0
2.0
1.4
0.4
1.6
1.8
1.2
–25 25 100 125 150
SHUTDOWN THRESHOLD
SHUTDOWN DELAY
S/S Pin Input Current
vs Voltage Error Amplifier Transconductance
vs Temperature
Switching Frequency
vs Feedback Pin Voltage
VC Pin Threshold and High
Clamp Voltage vs Temperature
FEEDBACK PIN VOLTAGE (V)
400
ERROR AMPLIFIER OUTPUT CURRENT (µA)
300
200
100
300
100
0.1 0.1
200
0
0.3 –0.2 V
REF
–55°C
125°C
25°C
LT1372 • G06
Minimum Synchronization
Voltage vs Temperature
TEMPERATURE (°C)
–50
0
MINIMUM SYNCHRONIZATION VOLTAGE (V
P-P
)
0.5
1.0
1.5
2.0
050
100 150
LT1372 • G05
2.5
3.0
–25 25 75 125
f
SYNC
= 700kHz (LT1372)
f
SYNC
= 1.4MHz (LT1377)
LT1377
LT1372
S/S PIN VOLTAGE (V)
–1
S/S PIN INPUT CURRENT (µA)
1
3
5
7
LT1372 • G07
–1
–3
0
2
4
–2
–4
–5 135
08
2469
V
IN
= 5V
FEEDBACK PIN VOLTAGE (V)
0
SWITCHING FREQUENCY (% OF TYPICAL)
70
90
110
0.8
LT1372 • G08
50
30
60
80
100
40
20
10 0.2 0.4 0.6
0.1 0.9
0.3 0.5 0.7 1.0
TEMPERATURE (°C)
–50
0
TRANSCONDUCTANCE (µmho)
200
600
800
1000
2000
1400
050 75
LT1372 • G09
400
1600
1800
1200
–25 25 100 125 150
g
m
= I (V
C
)
V (FB)
TEMPERATURE (°C)
–50
0.4
V
C
PIN VOLTAGE (V)
0.6
1.0
1.2
1.4
2.4
1.8
050 75
LT1372 • G10
0.8
2.0
2.2
1.6
–25 25 100 125 150
V
C
HIGH CLAMP
V
C
THRESHOLD
TEMPERATURE (°C)
–50
FEEDBACK INPUT CURRENT (nA)
400
500
600
150
LT1372 • G11
300
200
0050 100
100
800
700
–25 25 75 125
V
FB
=V
REF
Feedback Input Current
vs Temperature
TEMPERATURE (°C)
–50
–50
NEGATIVE FEEDBACK INPUT CURRENT (µA)
–30
0
050 75
LT1372 • G12
–40
–10
–20
–25 25 100 125 150
V
NFB
=V
NFR
Negative Feedback Input Current
vs Temperature
TYPICAL PERFOR A CE CHARACTERISTICS
UW
5
LT1372/LT1377
V
C
(Pin 1): The compensation pin is used for frequency
compensation, current limiting and soft start. It is the
output of the error amplifier and the input of the current
comparator. Loop frequency compensation can be per-
formed with an RC network connected from the V
C
pin to
ground.
FB (Pin 2): T
he feedback pin is used for positive output
voltage sensing and oscillator frequency shifting. It is the
inverting input to the error amplifier. The noninverting
input of this amplifier is internally tied to a 1.245V
reference. Load on the FB pin should not exceed 250µA
when the NFB pin is used. See Applications Information.
NFB (Pin 3): The negative feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 100k
source resistor.
S/S (Pin 4): Shutdown and Synchronization Pin. The S/S
pin is logic level compatible. Shutdown is active low and
the shutdown threshold is typically 1.3V. For normal
operation, pull the S/S pin high, tie it to V
IN
or leave it
floating. To synchronize switching, drive the S/S pin be-
tween 600kHz and 800kHz (LT1372) or 1.2MHz to 1.6MHz
(LT1377).
V
IN
(Pin 5): Bypass input supply pin with 10µF or more. The
part goes into undervoltage lockout when V
IN
drops below
2.5V. Undervoltage lockout stops switching and pulls the
V
C
pin low.
GND S (Pin 6): The ground sense pin is a “clean” ground.
The internal reference, error amplifier and negative feed-
back amplifier are referred to the ground sense pin. Con-
nect it to ground. Keep the ground path connection to the
output resistor divider and the V
C
compensation network
free of large ground currents.
GND (Pin 7): The ground pin is the emitter connection of
the power switch and has large currents flowing through it.
It should be connected directly to a good quality ground
plane.
V
SW
(Pin 8): The switch pin is the collector of the power
switch and has large currents flowing through it. Keep the
traces to the switching components as short as possible to
minimize radiation and voltage spikes.
+
NFBA
NFB
S/S
FB
100k
50k
0.08
+
EA
VC
VIN
GND LT1372 • BD
GND SENSE
1.245V
REF
5:1 FREQUENCY
SHIFT
OSCSYNC
SHUTDOWN
DELAY AND RESET LOW DROPOUT
2.3V REG ANTI-SAT
LOGIC DRIVER
SW
SWITCH
+
IA
AV 6
COMP
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PI FU CTIO S
BLOCK DIAGRA
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LT1372/LT1377
The LT1372/LT1377 are current mode switchers. This
means that switch duty cycle is directly controlled by
switch current rather than by output voltage. Referring to
the block diagram, the switch is turned “On” at the start of
each oscillator cycle. It is turned “Off” when switch current
reaches a predetermined level. Control of output voltage is
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at mid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely vary-
ing input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator pro-
vides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
500kHz (LT1372) or 1MHz (LT1377) oscillator is the basic
clock for all internal timing. It turns “On” the output switch
via the logic and driver circuitry. Special adaptive anti-sat
circuitry detects onset of saturation in the power switch
and adjusts driver current instantaneously to limit switch
saturation. This minimizes driver dissipation and provides
very rapid turn-off of the switch.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce out-
put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases ten times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps pro-
tect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator fre-
quency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.
Unique error amplifier circuitry allows the LT1372/LT1377
to directly regulate negative output voltages. The negative
feedback amplifier’s 100k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at –2.49V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. Consult Linear Technology Market-
ing for units that can regulate down to –1.25V.
The error signal developed at the amplifier output is
brought out externally. This pin (V
C
) has three different
functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regula-
tor operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (g
m
) type, so this voltage can
be externally clamped for lowering current limit. Like-
wise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the V
C
pin is pulled
below the control pin threshold, placing the LT1372/
LT1377 in an idle mode.
Positive Output Voltage Setting
The LT1372/LT1377 develops a 1.245V reference (V
REF
)
from the FB pin to ground. Output voltage is set by
connecting the FB pin to an output resistor divider
(Figure 1). The FB pin bias current represents a small
error and can usually be ignored for values of R2 up to 7k.
The suggested value for R2 is 6.19k. The NFB pin is
normally left open for positive output applications.
Figure 1. Positive Output Resistor Divider
R1 V
OUT
= V
REF
1 +
R2
FB
PIN
V
REF
V
OUT
()
R1
R2
R1 = R2 – 1
()
V
OUT
1.245
LT1372 • F01
OPERATIO
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APPLICATIO S I FOR ATIO
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LT1372/LT1377
Positive fixed voltage versions are available (consult
Linear Technology marketing).
Negative Output Voltage Setting
The LT1372/LT1377 develops a –2.49V reference (V
NFR
)
from the NFB pin to ground. Output voltage is set by
connecting the NFB pin to an output resistor divider
(Figure 2). The –30µA NFB pin bias current (I
NFB
) can
cause output voltage errors and should not be ignored.
This has been accounted for in the formula in Figure 2. The
suggested value for R2 is 2.49k. The FB pin is normally left
open for negative output application. See Dual Polarity
Output Voltage Sensing for limitatins on FB pin loading
when using the NFB pin.
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high, tied to V
IN
or left floating for normal operation.
A logic low on the S/S pin activates shutdown, reducing
the part’s supply current to 12µA. Typical synchronization
range is from 1.05 to 1.8 times the part’s natural switching
frequency, but is only guaranteed between 600kHz and
800kHz (LT1372) or 1.2MHz and 1.6MHz (LT1377). At
start-up, the synchronization signal should not be applied
until the feedback pin is above the frequency shift voltage
of 0.7V. If the NFB pin is used, synchronization should not
be applied until the NFB pin is more negative than –1.4V.
A 12µs resetable shutdown delay network guarantees the
part will not go into shutdown while receiving a synchro-
nization signal.
Caution should be used when synchronizing above 700kHz
(LT1372) or 1.4MHz (LT1377) because at higher sync
frequencies the amplitude of the internal slope compensa-
tion used to prevent subharmonic switching is reduced.
This type of subharmonic switching only occurs when the
duty cycle of the switch is above 50%. Higher inductor
values will tend to eliminate problems.
Thermal Considerations
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces-
sive die temperatures. The packages are rated at 120°C/W
for SO (S8) and 130°C/W for PDIP (N8).
Average supply current (including driver current) is:
I
IN
= 4mA + DC (I
SW
/60 + I
SW
× 0.004)
I
SW
= switch current
DC = switch duty cycle
Switch power dissipation is given by:
P
SW
= (I
SW
)
2
× R
SW
× DC
R
SW
= output switch “On” resistance
Total power dissipation of the die is the sum of supply
current times supply voltage plus switch power:
P
D(TOTAL)
= (I
IN
× V
IN
) + P
SW
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the “Dual
Output Flyback Converter with Overvoltage Protection”
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as de-
scribed above. When both the FB and NFB pins are used,
the LT1372/LT1377 acts to prevent either output from
going beyond its set output voltage. For example in this
application, if the positive output were more heavily loaded
than the negative, the negative output would be greater
and would regulate at the desired set-point voltage. The
positive output would sag slightly below its set-point
voltage. This technique prevents either output from going
unregulated high at no load. Please note that the load on
the FB pin should not exceed 250µA when the NFB pin is
used. This situation occurs when the resistor dividers are
used at
both
FB and NFB. True load on FB is not the full
divider current unless the positive output is shorted to
ground. See Dual Output Flyback Converter application.
R1 –V
OUT
= V
NFB
+ I
NFB
(R1)1 +
R2
LT1372 • F02
NFB
PIN
V
NFR
I
NFB
–V
OUT
()
R1
R2
R1 =
+ 30 × 10
6
V
OUT
– 2.49
( ) ( )
2.49
R2
Figure 2. Negative Output Resistor Divider
APPLICATIO S I FOR ATIO
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LT1372/LT1377
Choosing the Inductor
For most applications the inductor will fall in the range of
2.2µH to 22µH. Lower values are chosen to reduce physi-
cal size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 1.5A limit. Higher values also
reduce input ripple voltage and reduce core loss.
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation, and of course, cost.
The following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.
1. Assume that the average inductor current for a boost
converter is equal to load current times V
OUT
/V
IN
and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 0.5A, for instance,
a 0.5A inductor may not survive a continuous 1.5A
overload condition. Also be aware that boost convert-
ers are not short circuit protected, and that under
output short conditions, inductor current is limited only
by the available current of the input supply.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe-
cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving be-
cause they saturate softly, whereas ferrite cores satu-
rate abruptly. Other core materials fall in between
somewhere. The following formula assumes continu-
ous mode operation but it errors only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
IPEAK = IOUT ×
VIN = Minimum Input Voltage
f = 500kHz Switching Frequency (LT1372) or
1MHz Switching Frequency (LT1377)
+
VOUT
VIN
VIN(VOUT VIN)
2(f)(L)(VOUT)
3. Decide if the design can tolerate an “open” core geom-
etry like a rod or barrel, which have high magnetic field
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small, and there are no helpful
guidelines to calculate when the magnetic field radia-
tion will be a problem.
4. Start shopping for an inductor which meets the re-
quirements of core shape, peak current (to avoid
saturation), average current (to limit heating) and fault
current. If the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts. Keep in mind that
all good things like high efficiency, low profile and high
temperature operation will increase cost, sometimes
dramatically.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology application
department if you feel uncertain about the final choice.
They have experience with a wide range of inductor
types and can tell you about the latest developments in
low profile, surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its effective
series resistance, (ESR), because this is what determines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes
volume
, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1372 and LT1377 applications is
0.05 to 0.5. A typical output capacitor is an AVX type
TPS, 22µF at 25V, with a guaranteed ESR less than 0.2.
This is a “D” size surface mount solid tantalum capacitor.
TPS capacitors are specially constructed and tested for
low ESR, so they give the lowest ESR for a given volume.
To further reduce ESR, multiple output capacitors can be
used in parallel. The value in microfarads is not particu-
larly critical, and values from 22µF to greater than 500µF
work well, but you cannot cheat mother nature on ESR.
If you find a tiny 22µF solid tantalum capacitor, it will have
high ESR, and output ripple voltage will be terrible. Table
1 shows some typical solid tantalum surface mount
capacitors.
APPLICATIO S I FOR ATIO
WUUU
9
LT1372/LT1377
I
RIPPLE
=
f = 500kHz Switching frequency (LT1372) or,
1MHz Switching frequency (LT1377)
0.3(V
IN
)(V
OUT
– V
IN
)
(f)(L)(V
OUT
)
The input capacitor can see a very high surge current when
a battery or high capacitance source is connected “live”
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series, for instance), but even these units may
fail if the input voltage approaches the maximum voltage
rating of the capacitor. AVX recommends derating capaci-
tor voltage by 2:1 for high surge applications. Ceramic and
aluminum electrolytic capacitors may also be used and
have a high tolerance to turn-on surges.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumen-
tal in giving acceptable loop phase margin. Ceramic ca-
pacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Linear Technology plans to issue a Design Note on the use
of ceramic capacitors in the near future.
Output Diode
The suggested output diode (D1) is a 1N5818 Schottky or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch off time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
E CASE SIZE ESR (MAX ) RIPPLE CURRENT (A)
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
AVX TAJ 0.7 to 0.9 0.4
D CASE SIZE
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
AVX TAJ 0.9 to 2.0 0.36 to 0.24
C CASE SIZE
AVX TPS 0.2 (Typ) 0.5 (Typ)
AVX TAJ 1.8 to 3.0 0.22 to 0.17
B CASE SIZE
AVX TAJ 2.5 to 10 0.16 to 0.08
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the
output
capacitor. Solid
tantalum capacitors fail during very high
turn-on
surges,
which do not occur at the output of regulators. High
discharge
surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
Output Capacitor Ripple Current (RMS)
IRIPPLE (RMS) = IOUT
= IOUT
VOUT VIN
VIN
DC
1 – DC
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular and
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10µF to
100µF with an ESR of 0.3 or less work well up to full 1.5A
switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for
boost converter is :
APPLICATIO S I FOR ATIO
WUUU
10
LT1372/LT1377
(magnetic) radiation is minimized by keeping output di-
ode, switch pin, and output bypass capacitor leads as
short as possible. E field radiation is kept low by minimiz-
ing the length and area of all traces connected to the switch
pin. A ground plane should always be used under the
switcher circuitry to prevent interplane coupling.
The high speed switching current path is shown schemati-
cally in Figure 3. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode, and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
LOAD
V
OUT
L1 SWITCH
NODE
LT1372 • F03
V
IN
HIGH
FREQUENCY
CIRCULATING
PATH
Figure 3
More Help
For more detailed information on switching regulator
circuits, please see Application Note 19. Linear Technol-
ogy also offers a computer software program, SwitcherCAD,
to assist in designing switching converters. In addition,
our applications department is always ready to lend a
helping hand.
Frequency Compensation
Loop frequency compensation is performed on the output
of the error amplifier (V
C
pin) with a series RC network.
The main pole is formed by the series capacitor and the
output impedance (500k) of the error amplifier. The
pole falls in the range of 2Hz to 20Hz. The series resistor
creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor,
typically one-tenth the size of the main compensation
capacitor, is sometimes used to reduce the switching
frequency ripple on the V
C
pin. V
C
pin ripple is caused by
output voltage ripple attenuated by the output divider and
multiplied by the error amplifier. Without the second
capacitor, V
C
pin ripple is:
V
C
Pin Ripple =
V
RIPPLE
= Output ripple (V
P–P
)
g
m
= Error amplifier transconductance
(1500µmho)
R
C
= Series resistor on V
C
pin
V
OUT
= DC output voltage
1.245(V
RIPPLE
)(g
m
)(R
C
)
(V
OUT
)
To prevent irregular switching, V
C
pin ripple should be
kept below 50mV
P–P
.
Worst-case V
C
pin ripple occurs at
maximum output load current and will also be increased
if poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the V
C
pin reduces
switching frequency ripple to only a few millivolts. A low
value for R
C
will also reduce V
C
pin ripple, but loop phase
margin may be inadequate.
Switch Node Considerations
For maximum efficiency, switch rise and fall time are
made as short as possible. To prevent radiation and high
frequency resonance problems, proper layout of the com-
ponents connected to the switch node is essential. B field
APPLICATIO S I FOR ATIO
WUUU
11
LT1372/LT1377
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of circuits as described herein will not infringe on existing patent rights.
TYPICAL APPLICATIONS N
U
Low Ripple 5V to –3V “Cuk”
Converter
LT1372/LT1377
V
IN
S/S
GND
GND S
V
SW
NFB
V
C
5
4
7
6
8
3
1
+
+
R4
2k R2
4.99k
1%
R1
1k
1%
C4
0.047µF
C6
0.1µF
V
OUT
–3V
250mA
LT1372 • TA05
V
IN
5V
C3
47µF
16V
C1
22µF
10V
C2
47µF
16V
C5
0.0047µF
41
3
L1*
2
D1**
SUMIDA CLS62-100L
MOTOROLA MBR0520LT3
PATENTS MAY APPLY
*
**
+
Dual Output Flyback Converter with Overvoltage ProtectionPositive-to-Negative Converter with Direct Feedback
LT1372/LT1377
V
IN
FB
V
C
V
IN
2.7V TO 13V
1
3
8
52
4
6, 7
*DALE LPE-4841-100MB (605) 665-9301
GND
NFB
LT1372 • TA04
V
SW
S/S
P6KE-20A
1N4148
MBRS140T3
MBRS140T3
C1
22µF
R2
1.21k
1%
R1
13k
1%
C2
0.047µF
C3
0.0047µFR3
2k
R5
2.49k
1%
R4
12.1k
1%
–V
OUT
–15V
V
OUT
15V
C4
47µF
C5
47µF
ON
OFF
2, 3
6, 7
5
T1*
4
8
1
++
+
LT1372/LT1377
V
IN
V
C
V
IN
2.7V TO 16V
1
3
8
5
4
6, 7
*COILTRONICS CTX10-2 (407) 241-7876
GND
NFB
LT1372 • TA03
V
SW
S/S
D2
P6KE-15A
D3
1N4148
D1
MBRS130LT3
C1
22µF
C2
0.047µF
C3
0.0047µFR1
2k
R3
2.49k
1%
R2
2.49k
1%
–V
OUT
–5V
C4
47µF
ON
OFF
V
IN
3V
5V
9V
I
OUT
0.3A
0.5A
0.75A
2
1
4
T1*
3
MAX I
OUT
++
D2
1N4148
Q2
1N5818
D1
1N4148
562*
20k
DIMMING
10k
330
10
12345
Q1
10µFC1
0.1µF
V
IN
4.5V
TO 30V
V
IN
V
SW
V
FB
V
C
GND
S/S
5
84
2
16, 7
LT1372/LT1377
2µF
0.1µF
L1
33µH
T1
LT1372 • TA06
C1 = WIMA MKP-20
L1 = COILCRAFT DT3316-333
T1 = COILTRONICS CTX 110609
* = 1% FILM RESISTOR
DO NOT SUBSTITUTE COMPONENTS
Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001
LAMP
C2
27pF
5mA MAX
2.2µF
2.7V TO
5.5V
22k
1N4148
OPTIONAL REMOTE
DIMMING
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-6400
ON
OFF
CCFL BACKLIGHT APPLICATION CIRCUITS
CONTAINED IN THIS DATA SHEET ARE
COVERED BY U.S. PATENT NUMBER 5408162
AND OTHER PATENTS PENDING
+
+
+
90% Efficient CCFL Supply
12
LT1372/LT1377
TYPICAL APPLICATIONS N
U
2 Li-Ion Cell to 5V SEPIC Converter
LT1372/LT1377
V
IN
GND
V
IN
4V TO 9V
1
2
8
5
4
6, 7
V
C
FB
LT1372 • TA07
V
SW
S/S
C1
33µF
20V
C4
0.047µFC5
0.0047µF
R1
2k
R3
6.19k
1%
R2
18.7k
1%
V
OUT
5V
C3
100µF
10V
ON
OFF
V
IN
4V
5V
7V
9V
I
OUT
0.45A
0.55A
0.65A
0.72A
L1A*
10µH
L1B*
10µH
C2
1µF
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E105ZY5U-C103-F
C3 = AVX TPSD107M010R0100
*SINGLE INDUCTOR WITH TWO WINDINGS
COILTRONICS CTX10-1
MAX I
OUT
MBRS130LT3
+
+
RELATED PARTS
PACKAGE DESCRIPTION
U
Dimensions in inches (millimeters) unless otherwise noted.
PART NUMBER DESCRIPTION COMMENTS
LT1370 High Efficiency DC/DC Converter 42V, 6A, 500kHz Switch
LT1767 1.5A, 1.25MHz Step-Down Switching Regulator 3V to 25V Input, V
REF
= 1.2V, Synchronizable Up to 2MHz, MSOP Package
LT1374 High Efficiency Step-Down Switching Regulator 25V, 4.5A, 500kHz Switch
LTC1735-1 High Efficiency Step-Down Controller with Power Good Output Fault Protection, 16-Pin SSOP and SO-8
LTC®3402 Single Cell, High Current (2A), Micropower, Synchronous V
IN
= 0.7V to 5V, Up to 95% Efficiency Synchronizable Oscillator
3MHz Step-Up DC/DC Converter from 100kHz to 3MHz
LINEAR TECHNOLOGY CORPORATION 1995
sn13727 13727fbs LT/TP 0401 2K REV B • PRINTED IN THE USA
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
FAX: (408) 434-0507
www.linear-tech.com
N8 1098
0.009 – 0.015
(0.229 – 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.325 +0.035
–0.015
+0.889
–0.381
8.255
()
0.100
(2.54)
BSC
0.065
(1.651)
TYP
0.045 – 0.065
(1.143 – 1.651)
0.130 ± 0.005
(3.302 ± 0.127)
0.020
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.125
(3.175)
MIN
12 34
8765
0.255 ± 0.015*
(6.477 ± 0.381)
0.400*
(10.160)
MAX
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
0.016 – 0.050
(0.406 – 1.270)
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 1298
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**