LT8302/LT8302-3
1
Rev. G
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TYPICAL APPLICATION
FEATURES DESCRIPTION
42VIN Micropower No-Opto
Isolated Flyback Converter
with 65V/3.6A Switch
The LT
®
8302/LT8302-3 is a monolithic micropower iso-
lated flyback converter. By sampling the isolated output
voltage directly from the primary-side flyback waveform,
the part requires no third winding or opto-isolator for
regulation. The output voltage is programmed with two
external resistors and a third optional temperature com-
pensation resistor. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
Low ripple Burst Mode operation maintains high efficiency
at light load while minimizing the output voltage ripple. A
3.6A, 65V DMOS power switch is integrated along with all
the high voltage circuitry and control logic into a thermally
enhanced 8-lead SO package.
The LT8302/LT8302-3 operates from an input voltage
range of 3V to 42V and delivers up to 18W of isolated
output power. The high level of integration and the use of
boundary and low ripple burst modes result in a simple to
use, low component count, and high efficiency application
solution for isolated power delivery.
3V to 32VIN/5VOUT Isolated Flyback Converter
APPLICATIONS
n 3V to 42V Input Voltage Range
n 3.6A, 65V Internal DMOS Power Switch
n Low Quiescent Current:
n 106µA in Sleep Mode
n 380µA in Active Mode
n Quasi-Resonant Boundary Mode Operation at
Heavy Load
n Low Ripple Burst Mode
®
Operation at Light Load
n Minimum Load < 0.5% (Typ) of Full Output
n No Transformer Third Winding or Opto-Isolator
Required for Output Voltage Regulation
n Accurate EN/UVLO Threshold and Hysteresis
n Internal Compensation and Soft-Start
n Temperature Compensation for Output Diode
n Output Short-Circuit Protection
n Thermally Enhanced 8-Lead SO Package
n AEC-Q100 Qualified for Automotive Applications
n Isolated Automotive, Industrial, Medical
PowerSupplies
n Isolated Auxiliary/Housekeeping Power Supplies
Efficiency vs Load Current
VIN
LT8302/LT8302-3
SW
9µH
VIN
3V TO 32V 3:1
1µH
RFB
RREF
EN/UVLO
470pF
10µF
1µF
220µF
10mA TO 1.1A (VIN = 5V)
10mA TO 2.0A (VIN = 12V)
10mA TO 2.9A (VIN = 24V)
VOUT–
39Ω
154k
•
•
115k 10k
8302 TA01a
GND
INTVCC
TC
VOUT+
5V
LOAD CURRENT (A)
0
EFFICIENCY (%)
80
85
90
1.5 2.5
8302 TA01b
75
70
0.5 1.0 2.0 3.0
65
60
VIN = 5V
VIN = 12V
VIN = 24V
All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. patents, including 5438499, 7463497, 7471522.
LT8302/LT8302-3
2
Rev. G
For more information www.analog.com
PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS
SW (Note 2) ..............................................................65V
VIN ............................................................................42V
EN/UVLO ....................................................................VIN
RFB ........................................................VIN – 0.5V to VIN
Current Into RFB ....................................................200µA
INTVCC, RREF, TC .........................................................4V
Operating Junction Temperature Range (Notes 3, 4)
LT8302E, LT8302E-3 ......................... –40°C to 125°C
LT8302I, LT8302I-3 ........................... –40°C to 125°C
LT8302J, LT8302J-3 .......................... –40°C to 150°C
LT8302H, LT8302H-3 ........................ –40°C to 150°C
LT8302MP ......................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec) ................... 300°C
(Note 1)
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8302ES8E#PBF LT8302ES8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302IS8E#PBF LT8302IS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302JS8E#PBF LT8302JS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C
LT8302HS8E#PBF LT8302HS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C
LT8302MPS8E#PBF LT8302MPS8E#TRPBF 8302 8-Lead Plastic SO –55°C to 150°C
LT8302ES8E-3#PBF LT8302ES8E-3#TRPBF 83023 8-Lead Plastic SO –40°C to 125°C
LT8302IS8E-3#PBF LT8302IS8E-3#TRPBF 83023 8-Lead Plastic SO –40°C to 125°C
LT8302JS8E-3#PBF LT8302JS8E-3#TRPBF 83023 8-Lead Plastic SO –40°C to 150°C
LT8302HS8E-3#PBF LT8302HS8E-3#TRPBF 83023 8-Lead Plastic SO –40°C to 150°C
AUTOMOTIVE PRODUCTS**
LT8302ES8E#WPBF LT8302ES8E#WTRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302IS8E#WPBF LT8302IS8E#WTRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302JS8E#WPBF LT8302JS8E#WTRPBF 8302 8-Lead Plastic SO –40°C to 150°C
LT8302HS8E#WPBF LT8302HS8E#WTRPBF 8302 8-Lead Plastic SO –40°C to 150°C
LT8302ES8E-3#WPBF LT8302ES8E-3#WTRPBF 83023 8-Lead Plastic SO –40°C to 125°C
LT8302IS8E-3#WPBF LT8302IS8E-3#WTRPBF 83023 8-Lead Plastic SO –40°C to 125°C
LT8302JS8E-3#WPBF LT8302JS8E-3#WTRPBF 83023 8-Lead Plastic SO –40°C to 150°C
LT8302HS8E-3#WPBF LT8302HS8E-3#WTRPBF 83023 8-Lead Plastic SO –40°C to 150°C
Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix.
**Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These
models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your
local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for
thesemodels.
ORDER INFORMATION
1
2
3
4
8
7
6
5
TOP VIEW
TC
RREF
RFB
SW
EN/UVLO
INTVCC
VIN
GND
S8E PACKAGE
8-LEAD PLASTIC SO
θJA = 33°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
9
GND
LT8302/LT8302-3
3
Rev. G
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ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT
VIN VIN Voltage Range l3 42 V
IQVIN Quiescent Current VEN/UVLO = 0.2V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
0.5
53
106
380
2 µA
µA
µA
µA
EN/UVLO Shutdown Threshold For Lowest Off IQl0.2 0.75 V
EN/UVLO Enable Threshold Falling (E, I, H, MP Grades) l1.178 1.214 1.250 V
EN/UVLO Enable Threshold Falling (J Grade Only) l1.160 1.214 1.268 V
EN/UVLO Enable Hysteresis 14 mV
IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.3
–0.1
0
2.5
0
0.1
2.7
0.1
µA
µA
µA
VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.85 3 3.1 V
IINTVCC INTVCC Current Limit VINTVCC = 2.8V 10 13 20 mA
INTVCC UVLO Threshold Falling 2.39 2.47 2.55 V
INTVCC UVLO Hysteresis 105 mV
(RFB – VIN) Voltage IRFB = 75µA to 125µA –50 50 mV
RREF Regulation Voltage l0.98 1.00 1.02 V
RREF Regulation Voltage Line Regulation 3V ≤ VIN ≤ 42V –0.01 0 0.01 %/V
VTC TC Pin Voltage 1.00 V
ITC TC Pin Current VTC = 1.2V (LT8302)
VTC = 1.2V (LT8302-3)
VTC = 0.8V
12
7
15
10
–200
18
13
µA
µA
µA
fMIN Minimum Switching Frequency 11.3 12 12.7 kHz
tON(MIN) Minimum Switch-On Time 160 ns
tOFF(MAX) Maximum Switch-Off Time Backup Timer 170 µs
ISW(MAX) Maximum Switch Current Limit 3.6 4.5 5.4 A
ISW(MIN) Minimum Switch Current Limit 0.70 0.87 1.04 A
RDS(ON) Switch On-Resistance ISW = 1.5A 80 mΩ
ILKG Switch Leakage Current VSW = 65V 0.1 0.5 µA
tSS Soft-Start Timer 11 ms
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 65V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 65V as shown
in Figure5.
Note 3: The LT8302E/LT8302E-3 is guaranteed to meet performance
specifications from 0°C to 125°C junction temperature. Specifications
over the –40°C to 125°C operating junction temperature range are
assured by design, characterization and correlation with statistical process
controls. The LT8302I/LT8302I-3 is guaranteed over the full –40°C to
125°C operating junction temperature range. The LT8302J/LT8302J-3
and LT8302H/LT8302H-3 are guaranteed over the full –40°C to 150°C
operating junction temperature range. The LT8302MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8302/LT8302-3 includes overtemperature protection that
is intended to protect the devices during momentary overload conditions.
Junction temperature will exceed 150°C when overtemperature protection
is active. Continuous operation above the specified maximum operating
junction temperature may impair device reliability.
LT8302/LT8302-3
4
Rev. G
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TYPICAL PERFORMANCE CHARACTERISTICS
Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms
VIN Shutdown Current
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
Output Load and Line Regulation Output Temperature Variation
Switching Frequency
vs Load Current
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50
OUTPUT VOLTAGE (V)
5.1
5.2
5.3
25 75
8302 G02
5.0
4.9
–25 0 50 100 150125
4.8
4.7
FRONT PAGE APPLICATION
VIN = 12V
IOUT = 1A
RTC = 115k
RTC = OPEN
VSW
20V/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
IOUT = 2A
8302 G04
VSW
20V/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
IOUT = 0.5A
8302 G05
VSW
20V/DIV
VOUT
50mV/DIV
20µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
IOUT = 10mA
8302 G06
VIN (V)
0
IQ (µA)
6
8
10
40
8302 G07
4
2
010 20 30 50
TJ = 150°C
TJ = 25°C
TJ = –55°C
VIN (V)
0
80
IQ (µA)
90
100
110
120
130
140
10 20 30 40
8302 G08
50
TJ = 150°C
TJ = –55°C
TJ = 25°C
VIN (V)
0
IQ (µA)
380
400
420
40
8302 G09
360
340
320 10 20 30 50
TJ = 150°C
TJ = –55°C
TJ = 25°C
LOAD CURRENT (A)
0 0.5
0
FREQUENCY (kHz)
200
500
1.0 2.0 2.5
8302 G03
100
400
300
1.5 3.0
VIN = 5V
VIN = 12V
VIN = 24V
FRONT PAGE APPLICATION
LOAD CURRENT (A)
0
OUTPUT VOLTAGE (V)
5.15
1.5
8302 G01
5.00
4.90
0.5 1.0 2.0
4.85
4.80
5.20
5.10
5.05
4.95
2.5 3.0
VIN = 5V
VIN = 12V
VIN = 24V
LT8302/LT8302-3
5
Rev. G
For more information www.analog.com
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC Voltage vs VIN INTVCC UVLO Threshold (RFB-VIN) Voltage
RREF Regulation Voltage RREF Line Regulation TC Pin Voltage
EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50
VEN/UVLO (V)
1.220
1.230
150
8302 G10
1.210
1.200 050 100
–25 25 75 125
1.240
1.215
1.225
1.205
1.235 RISING
FALLING
TEMPERATURE (°C)
–50
IHYST (µA)
3
4
5
25 75 150
8302 G11
2
1
0
–25 0 50 100 125
TEMPERATURE (°C)
–50
2.80
VINTVCC (V)
2.85
2.90
2.95
3.00
0 50 100 150
8302 G12
3.05
3.10
–25 25 75 125
IINTVCC = 0mA
IINTVCC = 10mA
VIN (V)
5
VINTVCC (V)
2.95
3.00
3.05
35 4020 25 30
8302 G13
2.90
2.85
10 15 45
2.80
3.10
IINTVCC = 0mA
IINTVCC = 10mA
TEMPERATURE (°C)
–50
2.2
VINTVCC (V)
2.3
2.4
2.5
2.6
0 50 100 150
8302 G14
2.7
2.8
–25 25 75 125
FALLING
RISING
TEMPERATURE (°C)
–50
VOLTAGE (mV)
0
20
150
8302 G15
–20
–40 050 100
–25 25 75 125
40
–10
10
–30
30 IRFB = 125µA
IRFB = 100µA
IRFB = 75µA
TEMPERATURE (°C)
–50
0.990
VRREF (V)
0.992
0.996
0.998
1.000
1.010
1.004
050 75
8302 G16
0.994
1.006
1.008
1.002
–25 25 100 125 150
VIN (V)
0
VRREF (V)
1.002
1.006
1.010
40
8302 G17
0.998
0.994
1.000
1.004
1.008
0.996
0.992
0.990 10 20 30 50
TEMPERATURE (°C)
–50
VTC (V)
1.1
1.3
150
8302 G18
0.9
0.7 050 100
–25 25 75 125
1.5
1.0
1.2
0.8
1.4
LT8302/LT8302-3
6
Rev. G
For more information www.analog.com
Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time
RDS(ON) Switch Current Limit Maximum Switching Frequency
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50
RESISTANCE (mΩ)
120
160
200
25 75 150
8302 G19
80
40
0
–25 0 50 100 125
TEMPERATURE (°C)
–50
ISW (A)
3
4
5
25 75 150
8302 G20
2
1
0
–25 0 50 100 125
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
TEMPERATURE (°C)
–50
FREQUENCY (kHz)
300
400
500
25 75 150
8302 G21
200
100
0
–25 0 50 100 125
TEMPERATURE (°C)
–50
FREQUENCY (kHz)
12
16
20
25 75 150
8302 G22
8
4
0
–25 0 50 100 125
TEMPERATURE (°C)
–50
0
TIME (ns)
100
200
300
400
–25 0 25 50
8302 G23
75 100 125 150
TEMPERATURE (°C)
–50
0
TIME (ns)
100
200
300
400
–25 0 25 50
8302 G24
75 100 125 150
LT8302/LT8302-3
7
Rev. G
For more information www.analog.com
PIN FUNCTIONS
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8302/LT8302-3.
Pull the pin below 0.3V to shut down the LT8302/LT8302-
3. This pin has an accurate 1.214V threshold and can
be used to program a VIN undervoltage lockout (UVLO)
threshold using a resistor divider from VIN to ground. A
2.5µA current hysteresis allows the programming of VIN
UVLO hysteresis. If neither function is used, tie this pin
directly to VIN.
INTVCC (Pin 2): Internal 3V Linear Regulator Output. The
INTVCC pin is supplied from VIN and powers the inter-
nal control circuitry and gate driver. Do not overdrive the
INTV
CC
pin with any external supply, such as a third wind-
ing supply. Locally bypass this pin to ground with a mini-
mum 1µF ceramic capacitor.
VIN (Pin 3): Input Supply. The VIN pin supplies current to
the internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed
pad provides both electrical contact to ground and good
thermal contact to the printed circuit board. Solder the
exposed pad directly to the ground plane.
SW (Pin 5): Drain of the Internal DMOS Power Switch.
Minimize trace area at this pin to reduce EMI and voltage
spikes.
RFB (Pin 6): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer pri-
mary SW pin. The ratio of the RFB resistor to the RREF
resistor, times the internal voltage reference, determines
the output voltage (plus the effect of any non-unity trans-
former turns ratio). Minimize trace area at this pin.
RREF (Pin 7): Input Pin for External Ground Referred
Reference Resistor. The resistor at this pin should be in
the range of 10k, but for convenience in selecting a resis-
tor divider ratio, the value may range from 9.09k to 11.0k.
TC (Pin 8): Output Voltage Temperature Compensation.
The voltage at this pin is proportional to absolute tem-
perature (PTAT) with temperature coefficient equal to
3.35mV/°K, i.e., equal to 1V at room temperature 25°C.
The TC pin voltage can be used to estimate the LT8302/
LT8302-3 junction temperature. Connect a resistor from
this pin to the RREF pin to compensate the output diode
temperature coefficient.
LT8302/LT8302-3
8
Rev. G
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BLOCK DIAGRAM
–
+
–
+
3
2
8
DRIVER
INTVCC
VIN
T1
N:1
A2 RSENSE
A3
TC
8302 BD
RREF
RREF
REN2
REN1
RTC
RFB
–
+
gm
1.214V
1V
M4
OSCILLATOR
LDO
BOUNDARY
DETECTOR
START-UP,
REFERENCE,
CONTROL
PTAT
VOLTAGE
R M1
GND
4, EXPOSED PAD PIN 9
Q
S
M2
VIN
VIN
CIN
6
RFB
5
SW
L1A L1B COUT
DOUT
VOUT+
VOUT–
M3
25µA
INTVCC
1EN/UVLO
CINTVCC
2.5µA
1:4
7
–
+
A1
•
•
LT8302/LT8302-3
9
Rev. G
For more information www.analog.com
OPERATION
The LT8302/LT8302-3 is a current mode switching regula-
tor IC designed specially for the isolated flyback topology.
The key problem in isolated topologies is how to commu-
nicate the output voltage information from the isolated
secondary side of the transformer to the primary side
for regulation. Historically, opto-isolators or extra trans-
former windings communicate this information across
the isolation boundary. Opto-isolator circuits waste output
power, and the extra components increase the cost and
physical size of the power supply. Opto-isolators can also
cause system issues due to limited dynamic response,
nonlinearity, unit-to-unit variation and aging over life-
time. Circuits employing extra transformer windings also
exhibit deficiencies, as using an extra winding adds to
the transformer’s physical size and cost, and dynamic
response is often mediocre.
The LT8302/LT8302-3 samples the isolated output voltage
through the primary-side flyback pulse waveform. In this
manner, neither opto-isolator nor extra transformer wind-
ing is required for regulation. Since the LT8302/LT8302-3
operates in either boundary conduction mode or discon-
tinuous conduction mode, the output voltage is always
sampled on the SW pin when the secondary current is
zero. This method improves load regulation without the
need of external load compensation components.
The LT8302/LT8302-3 is a simple to use micropower iso-
lated flyback converter housed in a thermally enhanced
8-lead SO package. The output voltage is programmed
with two external resistors. An optional TC resistor
provides easy output diode temperature compensation.
By integrating the loop compensation and soft-start
inside, the part reduces the number of external compo-
nents. As shown in the Block Diagram, many of the blocks
are similar to those found in traditional switching reg-
ulators including reference, regulators, oscillator, logic,
current amplifier, current comparator, driver, and power
switch. The novel sections include a flyback pulse sense
circuit, a sample-and-hold error amplifier, and a boundary
mode detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Quasi-Resonant Boundary Mode Operation
The LT8302/LT8302-3 features quasi-resonant bound-
ary conduction mode operation at heavy load, where
the chip turns on the primary power switch when the
secondary current is zero and the SW rings to its valley.
Boundary conduction mode is a variable frequency, vari-
able peak-current switching scheme. The power switch
turns on and the transformer primary current increases
until an internally controlled peak current limit. After the
power switch turns off, the voltage on the SW pin rises to
the output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on at its valley.
LT8302/LT8302-3
10
Rev. G
For more information www.analog.com
OPERATION
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit subharmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode
increases the switching frequency and decreases the
switch peak current at the same ratio. Running at a higher
switching frequency up to several MHz increases switch-
ing and gate charge losses. To avoid this scenario, the
LT8302/LT8302-3 has an additional internal oscillator,
which clamps the maximum switching frequency to be
less than 380kHz. Once the switching frequency hits the
internal frequency clamp, the part starts to delay the switch
turn-on and operates in discontinuous conductionmode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8302/
LT8302-3 has to turn on and off at least for a minimum
amount of time and with a minimum frequency to allow
accurate sampling of the output voltage. The inherent
minimum switch current limit and minimum switch-off
time are necessary to guarantee the correct operation of
specific applications.
As the load gets very light, the LT8302/LT8302-3 starts to
fold back the switching frequency while keeping the min-
imum switch current limit. So the load current is able to
decrease while still allowing minimum switch-off time for
the sample-and-hold error amplifier. Meanwhile, the part
switches between sleep mode and active mode, thereby
reducing the effective quiescent current to improve light
load efficiency. In this condition, the LT8302/LT8302-3
runs in low ripple Burst Mode operation. The typical
12kHz minimum switching frequency determines how
often the output voltage is sampled and also the mini-
mum load requirement.
Difference Between LT8302 and LT8302-3
The difference between LT8302 and LT8302-3 is
theboundary detection method. The LT8302 is using the
dv/dt slope on RREF pin, while the LT8302-3 is using the
voltage level on R
REF
pin. For good transformers with
low leakage inductance, both the LT8302 and LT8302-3
are behaving the same. The LT8302-3 is recommended
for multiple-winding output applications due to its lower
sensitivity to the noise on RREF pin.
LT8302/LT8302-3
11
Rev. G
For more information www.analog.com
APPLICATIONS INFORMATION
Output Voltage
The RFB and RREF resistors as depicted in the Block
Diagram are external resistors used to program the out-
put voltage. The LT8302/LT8302-3 operates similar to
traditional current mode switchers, except in the use of a
unique flyback pulse sense circuit and a sample-and-hold
error amplifier, which sample and therefore regulate the
isolated output voltage from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current, IRFB,
by the RFB resistor and the flyback pulse sense circuit
(M2 and M3). This current, IRFB, also flows through the
RREF resistor to generate a ground-referred voltage. The
resulting voltage feeds to the inverting input of the sam-
ple-and-hold error amplifier. Since the sample-and-hold
error amplifier samples the voltage when the secondary
current is zero, the (ISEC • ESR) term in the VFLBK equation
can be assumed to be zero.
The internal reference voltage, VREF, 1.00V, feeds to the
noninverting input of the sample-and-hold error amplifier.
The relatively high gain in the overall loop causes the
voltage at the RREF pin to be nearly equal to the internal
reference voltage VREF. The resulting relationship between
VFLBK and VREF can be expressed as:
VFLBK
RFB
⎛
⎝
⎜⎞
⎠
⎟•RREF =VREF or
VFLBK =VREF •RFB
RREF
⎛
⎝
⎜⎞
⎠
⎟
VREF = Internal reference voltage 1.00V
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB and RREF resistors,
transformer turns ratio, and diode forward voltage:
VOUT =VREF •RFB
RREF
⎛
⎝
⎜⎞
⎠
⎟•1
NPS
⎛
⎝
⎜⎞
⎠
⎟– VF
Output Temperature Compensation
The first term in the VOUT equation does not have tem-
perature dependence, but the output diode forward
voltage, VF, has a significant negative temperature coef-
ficient (–1mV/°C to –2mV/°C). Such a negative tem-
perature coefficient produces approximately 200mV to
300mV voltage variation on the output voltage across
temperature.
For higher voltage outputs, such as 12V and 24V, the
output diode temperature coefficient has a negligible
effect on the output voltage regulation. For lower voltage
outputs, such as 3.3V and 5V, however, the output diode
temperature coefficient does count for an extra 2% to 5%
output voltage regulation.
The LT8302/LT8302-3 junction temperature usually tracks
the output diode junction temperature to the first order.
To compensate the negative temperature coefficient of the
output diode, a resistor, RTC, connected between the TC
and RREF pins generates a proportional-to-absolute-tem-
perature (PTAT) current. The PTAT current is zero at 25°C,
flows into the RREF pin at hot temperature, and flows out
of the RREF pin at cold temperature. With the RTC resistor
in place, the output voltage equation is revised as follows:
VOUT =VREF •RFB
RREF
•1
NPS
– VFTO
( )
– VTC / T
( )
•
T –TO
( )
•RFB
RTC
•1
NPS
– VF/ T
( )
•T–TO
( )
TO=Room temperature 25°
°
C
VF/ T
( )
=Output diode forward voltage
temperature coefficient
VTC / T
( )
=3.35mV/ C
LT8302/LT8302-3
12
Rev. G
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APPLICATIONS INFORMATION
To cancel the output diode temperature coefficient, the
following two equations should be satisfied:
VOUT =VREF •RFB
RREF
•1
NPS
– VFTO
( )
VTC/ T
( )
•RFB
RTC
•1
NPS
=– VF/ T
( )
Selecting Actual RREF, RFB, RTC Resistor Values
The LT8302/LT8302-3 uses a unique sampling scheme
to regulate the isolated output voltage. Due to the sam-
pling nature, the scheme contains repeatable delays and
error sources, which will affect the output voltage and
force a re-evaluation of the RFB and RTC resistor values.
Therefore, a simple 2-step sequential process is recom-
mended for selecting resistor values.
Rearrangement of the expression for V
OUT
in the previous
sections yields the starting value for RFB:
RFB =RREF •NPS •VOUT +VFTO
V
VOUT = Output voltage
V
F
(TO) = Output diode forward voltage at 25°C = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
The equation shows that the RFB resistor value is indepen-
dent of the R
TC
resistor value. Any R
TC
resistor connected
between the TC and RREF pins has no effect on the output
voltage setting at 25°C because the TC pin voltage is equal
to the RREF regulation voltage at 25°C.
The RREF resistor value should be approximately 10k
because the LT8302/LT8302-3 is trimmed and specified
using this value. If the RREF resistor value varies consid-
erably from 10k, additional errors will result. However, a
variation in RREF up to 10% is acceptable. This yields a
bit of freedom in selecting standard 1% resistor values
to yield nominal RFB/RREF ratios.
First, build and power up the application with the starting
R
REF
, R
FB
values (no R
TC
resistor yet) and other com-
ponents connected, and measure the regulated output
voltage, VOUT(MEAS). The new RFB value can be adjusted
to:
RFB(NEW) =
V
OUT
VOUT(MEAS)
•RFB
Second, with a new RFB resistor value selected, the output
diode temperature coefficient in the application can be
tested to determine the RTC value. Still without the RTC
resistor, the VOUT should be measured over temperature
at a desired target output load. It is very important for this
evaluation that uniform temperature be applied to both the
output diode and the LT8302/LT8302-3. If freeze spray or
a heat gun is used, there can be a significant mismatch
in temperature between the two devices that causes sig-
nificant error. Attempting to extrapolate the data from a
diode data sheet is another option if there is no method
to apply uniform heating or cooling such as an oven. With
at least two data points spreading across the operating
temperature range, the output diode temperature coeffi-
cient can be determined by:
–δVF/δT
( )
=VOUT T1
( )
– VOUT T2
( )
T1– T2
Using the measured output diode temperature coefficient,
an exact RTC value can be selected with the following
equation:
RTC =δVTC/δT
( )
–δVF/δT
( )
•RFB
NPS
⎛
⎝
⎜⎞
⎠
⎟
Once the RREF, RFB, and RTC values are selected, the reg-
ulation accuracy from board to board for a given appli-
cation will be very consistent, typically under ±5% when
including device variation of all the components in the
system (assuming resistor tolerances and transformer
windings matching within ±1%). However, if the trans-
former or the output diode is changed, or the layout is
dramatically altered, there may be some change in VOUT.
LT8302/LT8302-3
13
Rev. G
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APPLICATIONS INFORMATION
Output Power
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum out-
put current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and out-
put currents which make it similar to a nonisolated buck-
boost converter. The duty cycle will affect the input and
output currents, making it hard to predict output power. In
addition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
The graphs in Figure1 to Figure4 show the typical maxi-
mum output power possible for the output voltages 3.3V,
5V, 12V, and 24V. The maximum output power curve is
the calculated output power if the switch voltage is 50V
during the switch-off time. 15V of margin is left for leak-
age inductance voltage spike. To achieve this power level
at a given input, a winding ratio value must be calculated
to stress the switch to 50V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 8V and a maximum input
voltage of 32V. A three-to-one winding ratio fits this
design example perfectly and outputs equal to 15.3W at
32V but lowers to 7.7W at 8V.
Figure1. Output Power for 3.3V Output
Figure2. Output Power for 5V Output
Figure3. Output Power for 12V Output
Figure4. Output Power for 24V Output
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F02
5
010 20 30
MAXIMUM
OUTPUT POWER
20
N = 3:1
N = 1:1
N = 4:1
N = 2:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F03
5
010 20 30
N = 1:1
MAXIMUM
OUTPUT POWER
20
N = 3:2
N = 1:2
N = 2:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F04
5
010 20 30
N = 1:2
MAXIMUM
OUTPUT POWER
20
N = 2:3
N = 1:3
N = 1:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F01
5
010 20 30
N = 6:1
MAXIMUM
OUTPUT POWER
20
N = 4:1
N = 2:1
N = 3:1
LT8302/LT8302-3
14
Rev. G
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APPLICATIONS INFORMATION
The equations below calculate output power:
POUT = η • VIN • D • ISW(MAX) • 0.5
η = Efficiency = ~85%
D=Duty Cycle=VOUT +VF
( )
•NPS
VOUT +VF
( )
•NPS +VIN
I
SW(MAX)
= Maximum switch current limit = 3.6A (MIN)
Primary Inductance Requirement
The LT8302/LT8302-3 obtains output voltage information
from the reflected output voltage on the SW pin. The con-
duction of secondary current reflects the output voltage on
the primary SW pin. The sample-and-hold error amplifier
needs a minimum 350ns to settle and sample the reflected
output voltage. In order to ensure proper sampling, the
secondary winding needs to conduct current for a mini-
mum of 350ns. The following equation gives the minimum
value for primary-side magnetizing inductance:
LPRI ≥tOFF(MIN) •NPS • VOUT +VF
( )
ISW(MIN)
tOFF(MIN) = Minimum switch-off time = 350ns (TYP)
I
SW(MIN)
= Minimum switch current limit = 0.87A (TYP)
In addition to the primary inductance requirement for the
minimum switch-off time, the LT8302/LT8302-3 has mini-
mum switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
LPRI ≥
t
ON(MIN)
• V
IN(MAX)
ISW(MIN)
tON(MIN) = Minimum switch-on time = 160ns (TYP)
In general, choose a transformer with its primary mag-
netizing inductance about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8302/LT8302-
3. In addition to the usual list of guidelines dealing with
high frequency isolated power supply transformer design,
the following information should be carefully considered.
Analog Devices has worked with several leading magnetic
component manufacturers to produce pre-designed fly-
back transformers for use with the LT8302/LT8302-3.
Table1 shows the details of these transformers.
Table1. Predesigned Transformers–Typical Specifications
TRANSFORMER
PART NUMBER
DIMENSIONS
(W × L × H) (mm)
LPRI
(µH)
LLKG
(µH) NP:NS
RPRI
(mΩ)
RSEC
(mΩ) VENDOR
TARGET APPLICATION
VIN (V) VOUT (V) IOUT (A)
750311625 17.75 × 13.46 × 12.70 9 0.35 4:1 43 6 Wurth Elektronik 8 to 32 3.3 2.1
750311564 17.75 × 13.46 × 12.70 9 0.12 3:1 36 7 Wurth Elektronik 8 to 32 5 1.5
750313441 15.24 × 13.34 x 11.43 9 0.6 2:1 75 18 Wurth Elektronik 8 to 32 5 1.3
750311624 17.75 × 13.46 × 12.70 9 0.18 3:2 34 21 Wurth Elektronik 8 to 32 8 0.9
12387-TO79 15.5 × 12.5 × 11.5 9 0.5 1:1:1 55 90 Sumida 8 to 36 ±12 0.3
750313445 15.24 × 13.34 × 11.43 9 0.25 1:2 85 190 Wurth Elektronik 8 to 36 24 0.3
750313457 15.24 × 13.34 × 11.43 9 0.25 1:4 85 770 Wurth Elektronik 8 to 36 48 0.15
750313460 15.24 × 13.34 × 11.43 12 0.7 4:1 85 11 Wurth Elektronik 4 to 18 5 0.9
750311342 15.24 × 13.34 × 11.43 15 0.44 2:1 85 22 Wurth Elektronik 4 to 18 12 0.4
750313439 15.24 × 13.34 × 11.43 12 0.6 2:1 115 28 Wurth Elektronik 18 to 42 3.3 2.1
750313442 15.24 × 13.34 × 11.43 12 0.75 3:2 150 53 Wurth Elektronik 18 to 42 5 1.6
LT8302/LT8302-3
15
Rev. G
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APPLICATIONS INFORMATION
Turns Ratio
Note that when choosing an RFB/RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
3:1, 2:1, 1:1, etc., provides more freedom in settling total
turns and mutual inductance.
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V
or 5V), a N:1 turns ratio can be used with multiple pri-
mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (VLEAKAGE) on top of
this reflected voltage. This total quantity needs to remain
below the 65V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,
NPS, for a given application. Choose a turns ratio low
enough to ensure
NPS <
65V – V
IN(MAX)
– V
LEAKAGE
V
OUT
+V
F
For larger N:1 values, choose a transformer with a larger
physical size to deliver additional current. In addition,
choose a large enough inductance value to ensure that
the switch-off time is long enough to accurately sample
the output voltage.
For lower output power levels, choose a 1:1 or 1:N trans-
former for the absolute smallest transformer size. A 1:N
transformer will minimize the magnetizing inductance
(and minimize size), but will also limit the available output
power. A higher 1:N turns ratio makes it possible to have
very high output voltages without exceeding the break-
down voltage of the internal power switch.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer spec-
ifies turns ratio accuracy within ±1%.
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core
is saturated will not be transferred to the secondary and
will instead be dissipated in the core. When designing
custom transformers to be used with the LT8302/LT8302-
3, the saturation current should always be specified by
the transformer manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding
resistance due to the boundary/discontinuous conduction
mode operation of the LT8302/LT8302-3.
Leakage Inductance and Snubbers
Transformer leakage inductance on either the primary
or secondary causes a voltage spike to appear on the
primary after the power switch turns off. This spike is
increasingly prominent at higher load currents where
more stored energy must be dissipated. It is very import-
ant to minimize transformer leakage inductance.
When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in
Figure5, the reflected output voltage on the primary plus
VIN should be kept below 50V. This leaves at least 15V
margin for the leakage spike across line and load condi-
tions. A larger voltage margin will be required for poorly
wound transformers or for excessive leakage inductance.
LT8302/LT8302-3
16
Rev. G
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APPLICATIONS INFORMATION
tOFF > 350ns
VLEAKAGE
VSW
<65V
<50V
TIME 8302 F05
tSP < 250ns
Figure5. Maximum Voltages for SW Pin Flyback Waveform
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig-
ger boundary mode detector, the LT8302/LT8302-3 inter-
nally blanks the boundary mode detector for approximately
250ns. Any remaining voltage ringing after 250ns may turn
the power switch back on again before the secondary cur-
rent falls to zero. In this case, the LT8302/LT8302-3 enters
continuous conduction mode. So the leakage inductance
spike ringing should be limited to less than 250ns.
To clamp and damp the leakage voltage spikes, a
(RC + DZ) snubber circuit in Figure6 is recommended.
The RC (resistor-capacitor) snubber quickly damps the
voltage spike ringing and provides great load regulation
and EMI performance. And the DZ (diode-Zener) ensures
well defined and consistent clamping voltage to protect
SW pin from exceeding its 65V absolute maximum rating.
Figure6. (RC + DZ) Snubber Circuit
8302 F06
R
CZ
D
Lℓ
•
•
then add capacitance until the period of the ringing is 1.5
to 2 times longer. The change in period determines the
value of the parasitic capacitance, from which the para-
sitic inductance can be also determined from the initial
period. Once the value of the SW node capacitance and
inductance is known, a series resistor can be added to
the snubber capacitance to dissipate power and critically
damp the ringing. The equation for deriving the optimal
series resistance using the observed periods ( t
PERIOD
and
t
PERIOD(SNUBBED)
) and snubber capacitance (C
SNUBBER
) is:
CPAR =
C
SNUBBER
tPERIOD(SNUBBED)
tPERIOD
⎛
⎝
⎜⎞
⎠
⎟
2
–1
LPAR =tPERIOD2
CPAR • 4π2
RSNUBBER =LPAR
CPAR
Note that energy absorbed by the RC snubber will be con-
verted to heat and will not be delivered to the load. In high
voltage or high current applications, the snubber needs
to be sized for thermal dissipation. A 470pF capacitor in
series with a 39Ω resistor is a good starting point.
For the DZ snubber, proper care should be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage
inductance spike. Choose a diode that has a reverse-volt-
age rating higher than the maximum SW pin voltage. The
Zener diode breakdown voltage should be chosen to bal-
ance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown
with 5V margin. Use the following equation to make the
proper choice:
VZENNER(MAX) ≤ 60V – VIN(MAX)
For an application with a maximum input voltage of 32V,
choose a 24V Zener diode, the VZENER(MAX) of which is
around 26V and below the 28V maximum. The power loss
in the DZ snubber determines the power rating of the Zener
diode. A 1.5W Zener diode is typically recommended.
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin
when the power switch turns off without the snubber and
LT8302/LT8302-3
17
Rev. G
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APPLICATIONS INFORMATION
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin imple-
ments undervoltage lockout (UVLO). The EN/UVLO enable
falling threshold is set at 1.214V with 14mV hysteresis. In
addition, the EN/UVLO pin sinks 2.5µA when the voltage
on the pin is below 1.214V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
VIN(UVLO+)=
1.228V • R1
+
R2
( )
R2 +2.5µA •R
1
VIN(UVLO–)=1.214V • R1+R2
( )
R2
Figure7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8302/LT8302-3
GND
EN/UVLO
R1
RUN/STOP
CONTROL
(OPTIONAL)
R2
VIN
8302 F07
Figure7. Undervoltage Lockout (UVLO)
LT8302/LT8302-3 in shutdown with quiescent current
less than 2µA.
Minimum Load Requirement
The LT8302/LT8302-3 samples the isolated output
voltage from the primary-side flyback pulse waveform.
The flyback pulse occurs once the primary switch turns
off and the secondary winding conducts current. In order
to sample the output voltage, the LT8302/LT8302-3 has
to turn on and off for a minimum amount of time and with
a minimum frequency. The LT8302/LT8302-3 delivers a
minimum amount of energy even during light load con-
ditions to ensure accurate output voltage information.
The minimum energy delivery creates a minimum load
requirement, which can be approximately estimated as:
ILOAD(MIN) =
L
P
2
RI
•I
SW(MIN)
•f
MIN
2•V
OUT
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 1.04A (MAX)
fMIN = Minimum switching frequency = 12.7kHz (MAX)
The LT8302/LT8302-3 typically needs less than 0.5% of
its full output power as minimum load. Alternatively, a
Zener diode with its breakdown of 10% higher than the
output voltage can serve as a minimum load if pre-loading
is not acceptable. For a 5V output, use a 5.6V Zener with
cathode connected to the output.
Output Short Protection
When the output is heavily overloaded or shorted to
ground, the reflected SW pin waveform rings longer
than the internal blanking time. After the 350ns minimum
switch-off time, the excessive ringing falsely triggers the
boundary mode detector and turns the power switch back
on again before the secondary current falls to zero. Under
this condition, the LT8302/LT8302-3 runs into continu-
ous conduction mode at 380kHz maximum switching fre-
quency. If the sampled RREF voltage is still less than 0.6V
after 11ms (typ) soft-start timer, the LT8302/LT8302-3
initiates a new soft-start cycle. If the sampled RREF voltage
is larger than 0.6V after 11ms, the switch current may
run away and exceed the 4.5A maximum current limit.
Once the switch current hits 7.2A over current limit, the
LT8302/LT8302-3 also initiates a new soft-start cycle.
Under either condition, the new soft-start cycle throttles
back both the switch current limit and switch frequency.
The output short-circuit protection prevents the switch
current from running away and limits the average output
diode current.
LT8302/LT8302-3
18
Rev. G
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APPLICATIONS INFORMATION
Design Example
Use the following design example as a guide to design-
ing applications for the LT8302/LT8302-3. The design
example involves designing a 5V output with a 1.5A load
current and an input range from 8V to 32V.
VIN(MIN) = 8V, VIN(NOM) = 12V, VIN(MAX) = 32V,
VOUT = 5V, IOUT = 1.5A
Step 1: Select the transformer turns ratio.
NPS <
65V – V
IN(MAX)
– V
LEAKAGE
V
OUT
+V
F
VLEAKAGE = Margin for transformer leakage spike = 15V
VF = Output diode forward voltage = ~0.3V
Example:
NPS <
65V – 32V –15V
5V+0.3V
=3.4
The choice of transformer turns ratio is critical in deter-
mining output current capability of the converter. Table2
shows the switch voltage stress and output current capa-
bility at different transformer turns ratio.
Table2. Switch Voltage Stress and Output Current Capability vs
Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (A) DUTY CYCLE (%)
1:1 37.3 0.92 14-40
2:1 42.6 1.31 25-57
3:1 47.9 1.53 33-67
Clearly, only NPS = 3 can meet the 1.5A output current
requirement, so NPS = 3 is chosen as the turns ratio in
this example.
Step 2: Determine the primary inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI ≥tOFF(MIN) •NPS • VOUT +VF
( )
ISW(MIN)
LPRI ≥tON(MIN) • VIN(MAX)
ISW(MIN)
tOFF(MIN) = 350ns
tON(MIN) = 160ns
ISW(MIN) = 0.87A
Example:
LPRI ≥
350ns •3• 5V
+
0.3V
( )
0.87A =6.4µH
LPRI ≥160ns• 32V
0.87A
=5.9µH
Most transformers specify primary inductance with a tol-
erance of ±20%. With other component tolerance consid-
ered, choose a transformer with its primary inductance
40% to 60% larger than the minimum values calculated
above. LPRI = 9µH is then chosen in this example.
Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:
fSW =
1
tON +tOFF
=
1
LPRI •ISW
VIN
+LPRI •ISW
NPS • VOUT +VF
( )
ISW =VOUT •IOUT •2
η• V
IN
•D
LT8302/LT8302-3
19
Rev. G
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APPLICATIONS INFORMATION
Example:
D=5V
+
0.3V
( )
•3
5V +0.3V
( )
•3+12V =0.57
ISW =5V •1.5A •2
0.8 •12V •0.57
f
SW
=277kHz
The transformer also needs to be rated for the correct
saturation current level across line and load conditions.
A saturation current rating larger than 7A is necessary to
work with the LT8302/LT8302-3. The 750311564 from
Wurth is chosen as the flyback transformer.
Step 3: Choose the output diode.
Two main criteria for choosing the output diode include
forward current rating and reverse-voltage rating. The
maximum load requirement is a good first-order guess
at the average current requirement for the output diode.
Under output short-circuit condition, the output diode
needs to conduct much higher current. Therefore, a con-
servative metric is 60% of the maximum switch current
limit multiplied by the turns ratio:
IDIODE(MAX) = 0.6 • ISW(MAX) • NPS
Example:
IDIODE(MAX) = 8.1A
Next calculate reverse voltage requirement using maxi-
mum VIN:
VREVERSE =VOUT +
V
IN(MAX)
NPS
Example:
VREVERSE =5V+
32V
3
=15.7V
The PDS835L (8A, 35V diode) from Diodes Inc. is chosen.
Step 4: Choose the output capacitor.
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the following equation
to calculate the output capacitance:
COUT =
L
PRI
•I
SW
2
2• V
OUT
•ΔV
OUT
Example:
Design for output voltage ripple less than ±1% of VOUT,
i.e., 100mV.
COUT =9µH• 4.5A
( )
2
2•5V •0.1V
=182µF
Remember ceramic capacitors lose capacitance with
applied voltage. The capacitance can drop to 40% of
quoted capacitance at the maximum voltage rating. So
a 220µF, 6.3V rating X5R or X7R ceramic capacitor is
chosen.
Step 5: Design snubber circuit.
The snubber circuit protects the power switch from leak-
age inductance voltage spike. A (RC + DZ) snubber is
recommended for this application. A 470pF capacitor in
series with a 39Ω resistor is chosen as the RC snubber.
The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENNER(MAX) ≤ 60V – VIN(MAX)
Example:
VZENNER(MAX) ≤ 60V – 32V = 28V
A 24V Zener with a maximum of 26V will provide optimal
protection and minimize power loss. So a 24V, 1.5W Zener
from Central Semiconductor (CMZ5934B) is chosen.
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENNER(MAX)
Example:
VREVERSE > 60V
A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen.
LT8302/LT8302-3
20
Rev. G
For more information www.analog.com
APPLICATIONS INFORMATION
Step 6: Select the RREF and RFB resistors.
Use the following equation to calculate the starting values
for RREF and RFB:
RFB =RREF •NPS •VOUT +V
FTO
( )
( )
V
REF
R
REF
=10k
Example:
RFB =10k •3• 5V+0.3V
( )
1.00V
=159k
For 1% standard values, a 158k resistor is chosen.
Step 7: Adjust RFB resistor based on output voltage.
Build and power up the application with application com-
ponents and measure the regulated output voltage. Adjust
RFB resistor based on the measured output voltage:
RFB(NEW) =
V
OUT
VOUT(MEASURED)
•RFB
Example:
RFB =
5V
5.14V
•158k =154k
Step 8: Select RTC resistor based on output voltage tem-
perature variation.
Measure output voltage in a controlled temperature envi-
ronment like an oven to determine the output temperature
coefficient. Measure output voltage at a consistent load
current and input voltage, across the operating tempera-
ture range.
Calculate the temperature coefficient of VF:
–δVF/δT
( )
=VOUT T1
( )
– VOUT T2
( )
T1– T2
RTC =3.35mV/°C
–δVF/δT
( )
•RFB
NPS
⎛
⎝
⎜⎞
⎠
⎟
Example:
–δVF/δT
( )
=
5.189V – 5.041V
100°C– 0°C
( )
=1.48mV /°C
RTC =3.35mV/°C
1.48mV/°C•154
3
⎛
⎝
⎜⎞
⎠
⎟=115k
Step 9: Select the EN/UVLO resistors.
Determine the amount of hysteresis required and calcu-
late R1 resistor value:
VIN(HYS) = 2.5µA • R1
Example:
Choose 2V of hysteresis, R1 = 806k
Determine the UVLO thresholds and calculate R2 resistor
value:
V
IN(UVLO+)=
1.228V •R1+R2
( )
R2
+2.5µA •R
1
Example:
Set VIN UVLO rising threshold to 7.5V:
R2 = 232k
VIN(UVLO+) = 7.5V
VIN(UNLO–) = 5.5V
Step 10: Ensure minimum load.
The theoretical minimum load can be approximately esti-
mated as:
ILOAD(MIN) =9µH •1.04A
( )
2
•12.7kHz
2•5V
=12.4mA
Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con-
verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 10mA. In this example, a 500Ω resistor is selected
as the minimum load.
LT8302/LT8302-3
21
Rev. G
For more information www.analog.com
TYPICAL APPLICATIONS
8V to 32VIN/12VOUT Isolated Flyback Converter
8V to 32VIN/3.3VOUT Isolated Flyback Converter
Efficiency vs Load Current Load and Line Regulation
AMBIENT TEMPERATURE (°C)
–50
OUTPUT VOLTAGE (V)
3.30
3.40
150
8302 TA03b
3.20
3.10 050 100
–25 25 75 125
3.50
3.25
3.35
3.15
3.45
RTC = 105k
RTC = OPEN
VIN = 12V
IOUT = 1A
Output Temperature Variation
LOAD CURRENT (mA)
0
65
EFFICIENCY (%)
70
75
80
85
90
95
200 400 600 800 1000
8302 TA02b
1200
VIN = 12V
VIN = 24V
LOAD CURRENT (mA)
0
11.2
OUTPUT VOLTAGE (V)
11.4
11.6
11.8
12.0
12.2
12.4
200 400 600 800 1000 1200
8302 TA02c
VIN = 12V
VIN = 24V
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 32V
T1
1:1
9µH
RFB
RREF
C3
470pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
47µF
VOUT+
12V
5mA TO 0.8A (VIN = 12V)
5mA TO 1.1A (VIN = 24V)
VOUT–
R3
39Ω
R4
121k
•
•
R6
OPEN
R2
232k
R1
806k
R5
10k
D1: DIODES DFLS1100
D2: DIODES PDS360
T1: SUMIDA 12387-TO79
Z1: CENTRAL CMZ5934B
8302 TA02a
TC
EN/UVLO
GND
INTVCC
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 32V
T1
4:1
0.56µH
RFB
RREF
C3
470pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
470µF
VOUT+
3.3V
20mA TO 2.7A (VIN = 12V)
20mA TO 3.8A (VIN = 24V)
VOUT–
R3
39Ω
R4
140k
•
•
R6
105k
R2
232k
R1
806k
R5
10k
D1: DIODES DFLS1100
D2: DIODES PDS1040L
T1: WURTH 750311625
Z1: CENTRAL CMZ5934B
8302 TA03
TC
EN/UVLO
GND
INTVCC
LT8302/LT8302-3
22
Rev. G
For more information www.analog.com
TYPICAL APPLICATIONS
8V to 36VIN/±12VOUT Isolated Flyback Converter
8V to 36VIN/24VOUT Isolated Flyback Converter
8V to 36VIN/48VOUT Isolated Flyback Converter
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 36V
T1
1:1:1
9µH
RFB
RREF
C3
470pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
22µF
VOUT1+
12V
5mA TO 0.4A (VIN = 12V)
5mA TO 0.55A (VIN = 24V)
VOUT2–
R3
39Ω
R4
121k
•
•
R6
OPEN
R2
232k
R1
806k
R5
10k
D1: DIODES DFLS1100
D2, D3: DIODES PDS360
T1: SUMIDA 12387-TO79
Z1: CENTRAL CMZ5934B
8302 TA04
TC
9µH
D3
C5
22µF
VOUT2+
12V
5mA TO 0.4A (VIN = 12V)
5mA TO 0.55A (VIN = 24V)
VOUT2–
•
EN/UVLO
GND
INTVCC
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 36V
T1
1:2
36µH
RFB
RREF
C3
470pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
10µF
VOUT+
24V
2.5mA TO 0.4A (VIN = 12V)
2.5mA TO 0.55A (VIN = 24V)
VOUT–
R3
39Ω
R4
121k
•
•
R6
OPEN
R2
232k
R1
806k
R5
10k
D1: DIODES DFLS1100
D2: DIODES SBR2U150SA
T1: WURTH 750313445
Z1: CENTRAL CMZ5934B
8302 TA05
TC
EN/UVLO
GND
INTVCC
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 36V
T1
1:4
144µH
RFB
RREF
C3
470pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
2.2µF
VOUT+
48V
1.2mA TO 0.2A (VIN = 12V)
1.2mA TO 0.27A (VIN = 24V)
VOUT–
R3
39Ω
R4
121k
•
•
R6
OPEN
R2
232k
R1
806k
R5
10k
D1: DIODES DFLS1100
D2: DIODES SBR1U200P1
T1: WURTH 750313457
Z1: CENTRAL CMZ5934B
8302 TA06
TC
EN/UVLO
GND
INTVCC
LT8302/LT8302-3
23
Rev. G
For more information www.analog.com
TYPICAL APPLICATIONS
VIN
LT8302/LT8302-3
SW
9µH
VIN
8V TO 32V
T1
3:1
1µH
RFB
RREF
C3
470pF
Z1
D1 D2
C1
10µF
C2
1µF
C4
220µF
C4
10µF
C5
4.7µF
VOUT+
5V/2.0A (VIN = 12V)
5V/2.9A (VIN = 24V)
VOUT–
R3
39Ω
R7
5Ω
VCC
DRAIN
LT8309
GATE INTVCC
M1
GND
R4
154k
•
•
R6
OPEN
R2
232k
R1
806k
R8
2.1k
R5
10k
D1: DIODES DFLS1100
D2: CENTRAL CMMSH1-60
M1: INFINEON BSC059N04LS
T1: WURTH 750311564
Z1: CENTRAL CMZ5934B
8302 TA07
TC
EN/UVLO
GND
INTVCC
LOAD CURRENT (A)
0
EFFICIENCY (%)
85
90
95
1.5 2.5
8302 TA07b
80
75
0.5 1.0 2.0 3.0
70
65
8V to 32VIN/5VOUT Isolated Flyback Converter with LT8309
–4V to –42VIN/12VOUT Buck-Boost Converter
–18V to –42VIN/–12VOUT Negative Buck Converter
Efficiency vs Load Current
Efficiency vs Load Current
Efficiency vs Load Current
VIN SW
LT8302/LT8302-3
L1
12µH D1
Z1
GND
RFB
RREF
EN/UVLO
C3
47µF
D1: DIODES PMEG6030EP
L1: WURTH 744770112
Z1: CENTRAL CMHZ5243B
C2
1µF
VIN
–4V TO –42V
C1
10µF
R5
10k
8302 TA08a
VOUT
12V/0.45A (VIN = –5V)
12V/0.8A (VIN = –12V)
12V/1.1A (VIN = –24V)
12V/1.3A (VIN = –42V)
R4
118k
INTVCC
LOAD CURRENT (mA)
0
65
EFFICIENCY (%)
70
75
80
85
90
95
200 400 800600 1000 1200 1400
8302 TA08b
VIN = –5V
VIN = –12V
VIN = –24V
VIN = –42V
VIN
LT8302/LT8302-3
L1
12µH
VOUT
–12V
1.8A
D1 Z1
SW
RREF
EN/UVLO
RFB
EN/UVLO D1: DIODES PMEG6030EP
L1: WURTH 744770112
Z1: CENTRAL CMHZ5243B
C2
1µF
VIN
–18V TO –42V
C1
10µF
C3
47µF
R5
10k
8302 TA09a
R4
118k
R2
232k
R1
806k
INTVCC
LOAD CURRENT (mA)
0
70
EFFICIENCY (%)
75
80
85
90
95
100
500 1000 1500 2000
8302 TA09b
VIN = –18V
VIN = –24V
VIN = –42V
LT8302/LT8302-3
24
Rev. G
For more information www.analog.com
PACKAGE DESCRIPTION
.016 – .050
(0.406 – 1.270)
.010 – .020
(0.254 – 0.508)× 45°
0°– 8° TYP
.008 – .010
(0.203 – 0.254)
S8E 1015 REV C
.053 – .069
(1.346 – 1.752)
.014 – .019
(0.355 – 0.483)
TYP
.004 – .010
(0.101 – 0.254)
0.0 – 0.005
(0.0 – 0.130)
.080 – .099
(2.032 – 2.530)
.118 – .139
(2.997 – 3.550)
.050
(1.270)
BSC
1234
.150 – .157
(3.810 – 3.988)
NOTE 3
87
.005 (0.13) MAX
65
.189 – .197
(4.801 – 5.004)
NOTE 3
.228 – .244
(5.791 – 6.197)
.160 ±.005
(4.06 ±0.127)
.118
(2.99)
REF
RECOMMENDED SOLDER PAD LAYOUT
.045 ±.005
(1.143 ±0.127)
.050
(1.27)
BSC
INCHES
(MILLIMETERS)
NOTE:
1. DIMENSIONS IN
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm)
4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542)
5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542)
S8E Package
8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad
(Reference LTC DWG # 05-08-1857 Rev C)
.089
(2.26)
REF
.030 ±.005
(0.76
±0.127)
TYP
.245
(6.22)
MIN
45
LT8302/LT8302-3
25
Rev. G
For more information www.analog.com
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 11/14 Modified IQ and IHYS conditions.
Modified LPRI equation.
Modified schematic.
Updated Related Parts.
3
14
23
26
B 11/15 Revised package drawing. 24
C 9/16 Reduced EN/UVLO shutdown threshold.
Increased IINTVCC max current limit.
Changed ISW(MIN) current limit range.
Corrected ILOAD(MIN) equation.
3
3
3
20
D 5/19 Changed VIN minimum from 2.8V to 3V.
Table 1, Line 5: Replaced Wurth predesigned transformer with Sumida equivalent.
Table 1 Sumida transformer used in 12VOUT and ±12VOUT Typical Application circuits.
5V/1.1A (VIN = 5V) output capability line removed from LT8302/LT8302-3/LT8309 Typical Application circuit.
1, 3
14
21, 22
23
E 7/19 Added AEC-Q100 automotive models. 2
F 12/19 Added LT8302-3 Models All
G 04/20 Added J grade option and specifications 2, 3
LT8302/LT8302-3
26
Rev. G
For more information www.analog.com
ANALOG DEVICES, INC. 2013-2020
04/20
www.analog.com
RELATED PARTS
TYPICAL APPLICATION
4V to 42VIN/48VOUT Boost Converter
Efficiency vs Load Current
VIN SW
LT8302/LT8302-3
L1
22µH D1
Z1
GND
RFB
RREF
EN/UVLO C3
10µF
D1: DIODES PDS560
L1: WURTH 7443551221
Z1: CENTRAL CMHZ5262B
C2
1µF
VIN
4V TO 42V
C1
10µF
R5
10k
8302 TA10a
VOUT
48V/1.4A (VIN = 42V)
48V/0.8A (VIN = 24V)
48V/0.4A (VIN = 12V)
48V/0.15A (VIN = 5V)
R4
464k
R3
1M
INTVCC
LOAD CURRENT (mA)
0
70
EFFICIENCY (%)
75
80
85
90
100
250 500
8302 TA10b
1500750 1000 1250
95
VIN = 5V
VIN = 12V
VIN = 24V
VIN = 42V
PART NUMBER DESCRIPTION COMMENTS
LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A
Switch
Low IQ Monolithic No-Opto Flyback 5-Lead TSOT-23
LT8300 100VIN Micropower Isolated Flyback Converter with
150V/260mA Switch
Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23
LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23
LT3573/LT3574
LT3575
40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A
Switch
LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA
Switch, MSOP-16(12)
LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12)
LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components
LT3757A/LT3759
LT3758
40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch